Digital data transmission is becoming more and more important. In a digital data transmission system, a transmitted digital signal contains a sequence of encoded symbols each of which represents a predetermined number of data bits in the digital signal. One known method for coding such symbols is QAM, in which successive groups of bits (e.g. six or seven bits) are encoded into corresponding symbols. Each such symbol is represented by a complex signal, including an in-phase (or real) component I, and a quadrature (or imaginary) component Q. The value of this complex signal is one of a corresponding number (e.g. 64 or 128, respectively) of predetermined locations on the complex plane, called a constellation. This complex signal is then modulated onto the RF carrier. Other coding methods are known, including digital vestigial sideband (VSB) modulation, staggered QAM modulation, and quadrature phase shift keyed (QPSK) modulation. Digital signal receivers must be capable of receiving a digital signal, as described above, processing that signal, and reproducing the information represented by that signal, or storing that signal, e.g. on a magnetic tape, for reproduction at a later time. For example, television signals transmitted as a digital signal will soon supplement, and eventually replace, the analog television signals transmitted today. Television receivers will have to be able to receive digitally transmitted signals in any of the possible formats described above.
One aspect of digital receiver design is the synchronization of the receiver's sample clock to that of the transmitter; a process known as timing recovery. Several techniques exist for this synchronization. Conventional quadrature amplitude modulation (QAM) techniques include the use of a squaring loop. In this technique, the magnitude of the output signal from a sampling analog-to-digital converter is squared and bandpass filtered. This signal controls a phase locked loop, which in turn controls the phase and frequency of the receiver sampling clock for the analog-to-digital converter via a loop filter and voltage-controlled oscillator. This technique suffers deteriorating performance due to the presence of data noise in the loop. In addition, in the case of very dense constellations, the data modulation can completely mask the timing data and cause the synchronization circuit to fail to acquire and/or maintain synchronization.
Another synchronization technique is commonly known as band edge timing recovery. This technique is suitable for QAM constellations of any density, and with some modification may also be used with staggered QAM and VSB modulation. This technique operates independently of the QAM data carrier signal. Thus, timing may be recovered using this technique in the cresence of data carrier phase and frequency offsets. However, this technique tends to have a jitter component when it locks. To implement this technique for QAM requires only real filters, meaning that the filters processing the real and imaginary components operate independently of each other. However for VSB and staggered QAM, the filters required to implement this technique are analytic filters, meaning that they simultaneously process both real and imaginary components of a complex signal. The implementation of such filters requires four filter sections, which is a relatively complex implementation.
Synchronization techniques for QPSK modulated signals and for QAM modulated signals which are jitter-free are known. These techniques also operate independently of the QAM data carrier signal. In addition, these techniques use real filters which operate on the real and imaginary components independently, thus requiring only two filter sections. However, these techniques are not extensible to other forms of linear modulation, such as staggered QAM or VSB.
A timing recover technique is desirable which can provide jitter-free timing recovery, using real filters, and which can operate on signals using all forms of linear amplitude and/or phase modulation.
Another aspect of digital receiver design concerns the initial acquisition of a digital signal. This condition occurs whenever power is applied to the receiver, or when a new signal is tuned, e.g. the user changes the channel. When this condition occurs, the receiver must adapt to a new digital signal. One adaptation which must be made is the readjustment of the coefficients in the adaptive equalizer, which is used to compensate for channel characteristics, to a different channel path. Another adaptation which must be made is the recovery of the phase and frequency of the data carrier signal on which the digital data representing the digital signal is modulated (for example, the carrier of the QAM modulation, described above).
When the receiver is first turned on, or, for the example of a digital television receiver, when a new channel is selected by the viewer, the coefficients in the adaptive equalizer must be properly adjusted during the initial signal acquisition period. The adaptive equalizer compensates for channel characteristics and also suppresses intersymbol interference. In some digital transmission systems, a training sequence, i.e. a symbol sequence known to the receiver, is transmitted and the received signal is compared to the ideal signal representing the training sequence. On the basis of the received training sequence signal, the coefficients in the adaptive equalizer are adjusted so that the ideal training sequence signal is accurately reproduced at the receiver.
However, in other digital transmission systems, for example, in most digital television systems, it is not possible to use a training sequence to initially adjust the coefficients in the adaptive equalizer. Instead the actual received data signal must be used to adjust the adaptive equalizer coefficients, a process known as blind equalization. One known technique for adjusting the adaptive equalizer coefficients during the signal acquisition period is the constant modulus algorithm (CMA) technique. The CMA technique does not depend on decisions to operate, thus, accurate decisions are neither required, nor produced when the CMA technique is used.
The adaptive equalizer consists of two sections: a feedforward equalizer (FFE), which is coupled in series with the signal processing path; and a decision feedback equalizer (DFE), which normally processes decisions made by the quantizer, and is coupled in a feedback path from the output of the quantizer to a signal combiner at the output of the FFE. During the initial signal acquisition period, the CMA algorithm neither depends on, nor produces accurate decisions. Thus, two problems arise in adjusting the coefficients of the DFE during the signal acquisition period. First, what data should be supplied to the DFE during this period. Second, how can the coefficients of the DFE be adjusted when accurate decisions are not available to be fed to the DFE.
In one known method for adjusting the DFE coefficients, the FFE is rearranged to cover both its own time delay range and that of the DFE. This is done by moving the center tap of the FFE away from the end of the FFE, where it is normally set, toward the middle of the FFE. The coefficients of the FFE are then adjusted, using the error signal derived from the CMA technique (which, as described above, requires neither a recovered data carrier nor accurate decisions), until the coefficients of the FFE converge. At this point, the DFE is enabled, and decisions are supplied to its input terminal. The FFE tap coefficients in the time delay range normally covered by the DFE are migrated to the DFE taps by leaking the coefficients of the FFE taps in a known manner. The center tap of the FFE is then moved back toward the end of the FFE. This method is very complicated to control, and requires more complicated circuitry for the FFE and extra circuitry to migrate the taps from the FFE to the DFE.
Another problem with this method is that the FFE is not as effective at compensating for ISI as the DFE would be. Thus, it is possible that the FFE, operating alone, will not be able to compensate the channel sufficiently for the quantizer to begin its operation sufficiently accurately. It is therefore possible that even though the FFE has compensated the channel as best it can, when the DFE is actuated, the equalizer as a whole will still not be sufficiently converged for the coefficients of the DFE to converge.
An adaptive equalizer arrangement is desirable which permits convergence of the DFE coefficients during the initial signal acquisition period, which does not require the prior acquisition of the data carrier (i.e. can operate using the CMA algorithm), and which neither requires complicated control and filter circuitry, nor imposes an additional computational burden.
Once the equalizer coefficients have been converged, there are two basic types of techniques for acquiring and tracking the phase and frequency of the data subcarrier: decision directed and non-decision directed. In decision directed techniques, the output from the quantizer, which is the sequence of received symbols corresponding to an estimate of the transmitted symbol sequence, is used to control the process of acquiring and tracking the data carrier signal. Non-decision directed techniques do not use the quantizer output signal. It is known that decision directed techniques have a problem when a receiver begins operation, or, as in the case of a television receiver, begins to process a newly selected signal. Channel impairments remaining after the blind equalization process, and initial inaccuracy in the data carrier timing recovery, cause the initial quantizer decisions to be wrong a substantial proportion of the time. The incorrect decisions, then, will prevent the data carrier from being properly acquired.
In order to facilitate the acquisition of the data carrier, it is known to produce decisions based on an artificially coarse constellation consisting of four constellation points: one in each quadrant. This solution is based on the assumption that while a decision based on a full constellation is likely to be wrong a substantial proportion of the time during the data carrier acquisition period, the sign of the I and Q components are likely to be correct. Thus, a decision based only on the sign of the I and Q components will less likely be erroneous, and will enable acquisition of the data carrier. The quantizer is operated in this quadrant mode until the accuracy of the decisions exeeds a predetermined threshold level. At that point, the quantizer is switched into the full decision mode.
However, it is possible that, despite the accuracy of the quantizer decisions in the quadrant mode, the data carrier timing will not be sufficiently accurate to complete the acquisition of the data carrier when the quantizer switches into the full decision directed mode. In such a case, it can take a relatively long period to complete the acquisition of the data carrier after switching to the full decision directed mode. It is even possible that the data carrier will never be acquired, allowing entry into the full decision mode, because the best accuracy achieved by the quantizer in the quadrant mode is not sufficient to retain the lock when switching to the full decision directed mode.
A data carrier acquisition and tracking technique is desirable which can always successfully acquire the phase and frequency of the data carrier from initial tuning of an equalized digital television signal until the quantizer begins operation in the full decision directed mode.
Yet another aspect of digital television receiver design concerns coexistence with the existing analog television system. Future digital television systems will coexist with existing analog television systems, possibly in the same spectral space. The analog signal produced by existing analog systems will appear to the digital television signal as a narrowband and continuous wave interference (CWI) signal. To prevent degradation in the performance of the digital television signal, such CWI signals must be suppressed.
Known methods for suppressing CWI interference involve an adaptive notch filter, followed by a DFE. The notch of the filter is adjusted to the spectral location of the CWI, and the DFE compensates for the notch inserted by the adaptive notch filter. But, adaptive notch filters are complicated to implement. A CWI suppression system is desirable which can suppress CWI without requiring a complicated adaptive notch filter.