text: an embodiment of the invention is described with reference to the figures using reference designations as shown in the figures . referring to fig1 and 3 , the baseband time - domain basic measurement test configuration system consists of an upconverting transmitter 10 and a downconverting receiver 12 , both driven by a local oscillator generator 14 , for measuring baseband waveforms . the oscillator generator 14 includes a local oscillator ( lo ) 16 , a splitter 18 providing a local oscillator signal to an isolator 20 and phase shifter 22 . the isolator 20 provides a fixed phase local oscillator signal 24 that may be considered as a carrier signal . the phase shifter 22 provides a variable phase local oscillator signal 25 that is used for coherent downconversion . in a broad form , the preferred transmitter 10 is driven by a baseband signal source 26 for providing a baseband test signal to the transmitter 10 and for providing a 10 mhz reference signal 27 and a trigger signal 59 . the transmitter 10 includes an upconverter 28 , microwave amplifier 30 and attenuator 32 . the baseband signal source 26 provides the baseband test signal that drives the upconverter 28 receiving the fixed phase lo signal 24 for providing an upconverted signal to the microwave amplifier 30 that in turn drives the attenuator 32 providing a microwave test signal 34 . when testing a device under test ( dut ) 36 , the microwave test signal is applied directly to the dut 36 providing a microwave response signal 38 . when characterizing an arbitrary microwave test signal , such as the microwave test signal 34 , the microwave test signal 34 becomes a microwave test signal 38 applied directly to the receiver 12 . hence , the transmitter 10 can be any arbitrary system providing an arbitrary repetitive modulated microwave test signal 34 . in the broad form of the invention , the microwave test signal 34 is the microwave test signal 38 connected directly to the receiver 12 for characterization of the microwave test signal 34 and 38 without characterizing the dut 36 with the dut 36 effectively removed as a short circuit having no responsive characteristics . in preferred form , the dut 36 is inserted between the transmitter 10 and receiver 12 . the receiver 12 includes a downconverter 40 and a baseband amplifier 42 . when excited by the microwave test signal 34 , the inserted dut 36 provides a microwave response signal 38 to a downconverter 40 that also receives the variable phase lo signal 25 and provides a downconverted signal to the baseband amplifier 42 that in turn provides a baseband response signal that is recorded by an analyzer 44 . the microwave signals 34 and 38 of the dut to be measured can have any arbitrary repetitive phase or amplitude modulation , but the microwave signals 34 and 38 must be accompanied by unmodulated lo signals 24 and 25 to respective upconverter 28 and downconverter 40 of transmitter 10 and receiver 12 , respectively . the trigger signal 59 must also be supplied to analyzer 44 , while the reference signal 27 is optional . the transmitter 10 and receiver 12 operate respectively using the fixed phase lo signal 24 and the variable phase lo signal 25 , which are coherent lo signals . these coherent lo signals 24 and 25 are used during testing and calibration procedures . the analyzer 44 is preferably a microwave transition analyzer ( mta ), but could also be a digital storage oscilloscope ( dso ) used in place of the mta with reduced accuracy , because most dsos do not have the calibrated accuracy of an mta . referring to fig2 the transmitter 10 may further include an isolator 50 connected between the microwave amplifier 30 and attenuator 32 . referring to fig3 the downconverting receiver 12 may further include a bias tee coupler 54 for providing a voltage signal to a volt meter 56 . a power meter 57 is used to measure the power of the microwave test signal , from the transmitter 10 , or the microwave response signal 38 when the dut 36 is inserted . the receiver 12 may further include attenuators 58 and 60 for impedance matching . the dc level of the downconverted signal from the downconverter 40 is separated from the lpe signal by the bias tee 54 and corresponds to the fourier component at the lo frequency . the dc level that is separated by the bias tee 54 is measured by the volt meter 56 at the output of the downconverter 40 that may also be a mixer . the measurement of the dc level is preferred because the baseband amplifier 42 blocks the dc component of the baseband response signal from the downconverter 40 . the frequency stability requirements on the local oscillator signals 24 and 25 from oscillator 16 need not be high because the phase noise is canceled by downconversion in the receiver 12 . the baseband modulation signal provided to the upconverter 28 , which is a mixer 28 , however , must be stable , preferably with a 10 . 0 mhz reference output 27 to be used by the mta 44 as an external reference . the modulating baseband source 26 must also be used to trigger the mta 44 using trigger signal 59 . the method is used to characterize microwave test signals 34 and / or a dut 36 . firstly , the frequency response of the downconverting receiver 12 , including the downconverter 40 is determined . the downconverter 40 is a frequency translating device . the downconverting receiver frequency response is determined using the known frequency translation device characterization method of the related patent . secondly , a microwave test signal 34 is generated and applied directly to the downconverting receiver 12 . the downconverted baseband signal from the amplifier 42 is then recorded by the analyzer 44 , where the phase shifter , 22 , is at a known setting designated as 0 degrees to record the uncorrected in - phase component of the lpe signal of the baseband signal with the dut inserted . the phase shifter 22 is then adjusted by ninety degrees to a setting designated as ninety degrees and the downconverted baseband signal is again recorded to form the uncorrected quadrature component of the lpe signal . the in - phase and quadrature waveforms are recorded at the 0 ° and 90 ° phase settings respectively . in the preferred form , the dc voltage is also recorded at the same two phase settings by means of the volt meter 56 to produce the dc component of the lpe waveforms . alternatively , the dc measurement performed by means of the volt meter 56 may be recorded at four phase settings , 0 °, 90 °, 180 °, and 270 ° in order to more effectively cancel dc mixer offsets . in this instance , the dc component at 0 ° and the inverse of the dc component at 180 ° are combined to form the in - phase dc component , and the dc component at 90 ° and the inverse of the dc component at 270 ° are combined to form the quadrature dc component . the lpe signals from the amplifier 42 includes the frequency response of the entire downconverting receiver 12 and are therefore uncorrected , in view of the microwave test signal 34 . to obtain corrected lpe signals , the frequency response of the downconverting receiver 12 is removed . the correction procedure is required to remove the receiver frequency response from the baseband test signal . thirdly , the dut is inserted between the transmitter 10 and receiver 12 , and the microwave test signal 34 is applied to the dut providing the microwave response signal 38 to the downconverting receiver 12 . the microwave response signal 38 is measured in the identical way as the microwave test signal was measured as described immediately above . that is , the analyzer 44 records the downconverted , baseband in - phase and quadrature waveforms at the two phase shifter settings , 0 degrees and 90 degrees respectively , and the voltmeter 56 measures the dc component of the baseband signal at the same two phase shifter settings . these waveforms are then corrected for the downconverting receiver frequency response to obtain the corrected characterization of the microwave response signal 38 . the input signal 34 of the dut and the output signal 38 of the dut have thus been obtained so that the input - output characteristics of the dut is recorded . hence , the method is a three part method that includes firstly the characterization of the receiver 12 comprising the downconverter 40 and amplifier 42 , a characterization of the microwave test signal 34 and a characterization of the microwave response signal 38 . the characterization of the microwave test signal 34 and of the microwave response signal 38 together constitute the characterization of the input - output response of the dut . in the special case where the dut is linear , the frequency response of the dut may be determined directly from the uncorrected lpe test signal and the uncorrected lpe response signal . in this case , the corrected lpe test signal or the corrected lpe response signal need not be calculated . this is because the linear dut frequency response is equal to the discrete fourier transform of the uncorrected lpe response signal divided by the discrete fourier transform of the uncorrected lpe test signal . the transmitter 10 provides the microwave test signal 34 to the dut 36 or to the receiver 12 to produce the baseband lpe waveform that is recorded by the mta 44 . at the same time , the dc component at the phase shifter setting is recorded , scaled to take into account the gain of the baseband amplifier 42 , and added to the baseband lpe waveform . the phase shifter 22 is then adjusted by 90 ° and the downconverted baseband signal and the corresponding dc component are again recorded and combined to yield the uncorrected quadrature component of the lpe signal . enhanced measurement accuracy of the fourier component at dc is provided by performing a zeroing procedure consisting of another dc measurement at both phase settings with no microwave test signal input and subtracting the dc values thus obtained from the measurements with the microwave test signal applied . this zeroing procedure may be applied periodically to eliminate drift . if the ultimate dc component accuracy is required , measurements at four phase settings 90 ° apart may used to provide enhanced cancellation of dc mixer offsets . the 0 ° and the inverse of the 180 ° measurements are combined to form the in - phase component , and the 90 ° and the inverse of the 270 ° measurements are combined to form the quadrature component . this procedure eliminates any drift that can occur during the time interval between the dc zeroing procedure and the time at which the baseband waveform data is recorded . because the same downconverting receiver 12 is used to measure both the i and the q waveforms , there is no i / q imbalance . the lpe signals thus obtained include the frequency response of the receiver 12 and are therefore uncorrected . to obtain corrected lpe signals , the frequency response of the receiver 12 must be removed . the response of the receiver 12 is removed analytically from the uncorrected signals by means of the prior baseband - double - sideband frequency translating device ftd characterization method . the receiver response may then be removed analytically from the uncorrected lpe signal measurements , leaving an accurate lpe representation of the microwave test signal . among the signals that can be recorded are input microwave test signal 34 with the dut 36 removed , and the output microwave response signal 38 of a nonlinear device under test 36 when inserted and excited by the input microwave test signal 34 . the transmitter 10 and the receiver 12 are connected together to record the input lpe signal , that is the microwave test signal 34 , and then the microwave response signal 38 of the dut 36 when inserted between the transmitter 10 and the receiver 12 to record the microwave response signal 38 as an output lpe signal . the broad form of the method is to record the output waveforms of an arbitrary microwave frequency modulator 10 . these time - domain measurements can then be used to optimize and validate simulation models of arbitrary modulators 10 and / or duts 36 . to enhance the usefulness of the time - domain measurement system , the upconverting transmitter 10 is used with the coherent lo generator 16 and a baseband waveform source 26 to provide modulated microwave test signals 34 . the microwave amplifier 30 and variable attenuator 32 enable the input power to the dut 36 to be adjusted over an predetermined operating range . once the uncorrected baseband signals are measured with the dut 36 inserted between the transmitter 10 and receiver 12 , the receiver frequency response are analytically removed . although any modulated microwave test signal 34 may be used , for example , a 0 . 35 - ns wide , 0 . 5 v amplitude baseband pulse generated by source 26 may be used as a modulation input to the upconverter 28 . the lo frequency of signals 24 and 25 may be for this example , 19 . 6 ghz . the transmitted pulse can be measured using the baseband time - domain measurement method and the receiver response correction then applied . as a validation of the time - domain baseband method , the same microwave test signal 34 is measured directly at 19 . 6 ghz using the mta without the dut 36 or downconverting receiver 12 inserted . referring to all of the figures , and more particularly to fig4 a and 4 b , the signal envelope of both a corrected baseband measurement and a direct microwave measurement of a microwave pulse at 19 . 6 ghz are shown . the direct microwave measurement without the dut 36 or receiver 12 was recorded by analyzer 44 and then downconverted to baseband in software . the agreement between the corrected baseband and the direct microwave measurement indicates the validity of the time - domain baseband measurement method . this comparison illustrates the consistency of the time - domain baseband method with the direct measurement . the time - domain baseband method may be applied to situations where the direct microwave measurement is inadequate , for example where the lo frequency is beyond the range of the analyzer 44 , or the phase noise is too high , or the repetition period of the signal is too long to allow nyquist sampling of the signal at the lo frequency given the limited number of samples available for example , 1024 samples for the mta . measuring signals at baseband has several advantages compared to measuring them directly at the lo frequency . one advantage is that the sample rate can be reduced by the ratio of half the signal bandwidth to the lo frequency , thereby allowing for a longer time record or higher time resolution of the signal for the same number of samples . another advantage is that the time base accuracy and stability requirements for the analyzer 44 ( dso or mta ) are reduced by the same ratio of half the signal bandwidth to the carrier frequency . yet another advantage is that the phase noise of the carrier is eliminated because the same lo is used for both the upconversion and the downconversion . finally , the mta or dso has a flatter frequency response at baseband than at high lo frequencies . for example , signals with a 4 . 0 ghz bandwidth require only 2 . 0 ghz at baseband , over which the mta has a gain variation of only +/− 0 . 1 db . the phase deviation from linearity can be as high as 40 ° for a typical digitizing oscilloscope at 20 . 0 ghz , but it is negligible below 5 . 0 ghz . fig4 a and 4b show an example of time - domain measurements of a sub - nanosecond amplitude pulse . this waveform was generated by applying a 0 . 35 - ns width , + 0 . 2 - v amplitude baseband pulse to the upconverting transmitter shown in fig3 using an lo frequency of 19 . 6 ghz . this signal was measured by means of the baseband - time - domain measurement method where the waveform recording instrument was an mta . the same signal was also measured directly at the 19 . 6 ghz carrier frequency also using the mta with a sampling frequency of four samples per cycle of the carrier frequency . the prior art direct microwave measurement was then downconverted in software to baseband for comparison with the time - domain baseband measurement method . the in - phase and quadrature waveforms generated by the two methods are compared as shown in fig4 . the excellent agreement indicates the consistency of the two measurement techniques . the response of the mta has not been removed from the data shown . the increased amplitude and phase distortion of the mta in the direct microwave measurement may account for the slight differences between the results of the two methods . the baseband time domain measurement method preferably uses the ftd measurement method to first measure the response of the downconverting receiver 12 to then be able to remove the distortion of the downconverting receiver in baseband time - domain measurements . the use of the same downconverting mixer ftd 40 with a phase - shifted lo eliminates i / q imbalance in the measured complex baseband waveform . the four lo phases , 90 ° apart , are used to remove mixer dc offsets in the time - domain data , and used in calibration of the mixer offsets by measuring the dc output of the downconverter 40 with no signal input to achieve additional accuracy enhancement . referring to fig5 a - 5d , the time - domain baseband method preferably uses a detailed processing algorithm as a process for measuring the time - domain response of the dut in ten steps . step 1 measures the transmission response of the receiver 12 by means of the single - sideband frequency translating device measurement method or the double - sideband frequency translating device measurement method as taught in the related patent . step 1 provides the lowpass equivalent response of the downconverter 40 . this method has , as an output , the transmission response of the receiver h meas ( ω meas ( m )). this transmission response will be used in subsequent steps to remove the receiver distortion from the time domain baseband waveforms . steps 2 and 3 are required to obtain accurate calibrated measurements of a fourier component of the signals at the local oscillator generator frequency . this fourier component is at the local oscillator frequency at the microwave signal input 38 to the receiver 12 . this fourier component at the local oscillator generator frequency is downconverted to baseband by the mixer 40 and the downconverted baseband signal is provided at the intermediate frequency output of the mixer 40 . the fourier signal component appears as a dc voltage . the dc voltage is separated from the other signal components of the baseband signal by the bias tee 54 , and this dc voltage is then measured by the voltmeter 56 . step 2 measures the dc voltage by means of the voltmeter 56 , with no signal input to the receiver at input 38 . the dc voltage is measured and recorded at two settings of the phase shifter 22 . the two phase shifter settings must be 90 degrees apart at the local oscillator frequency . io is the voltage at the 0 ° setting and qo is the voltage at the 90 degree setting . step 3 measures the dc voltage by means of the voltmeter 56 , with the fixed phase lo signal 24 considered a continuous wave ( cw ) signal 24 used as a calibration signal 24 from the local oscillator generator 16 , and injected directly into the receiver at input 38 . the power in this continuous wave signal 24 and 38 is measured by means of a power meter 57 , and recorded as pcw . the dc voltage is measured and recorded at two settings of the phase shifter 22 . the two phase shifter settings are 90 degrees apart at the local oscillator frequency , and are the same settings used in step 2 . icw is the voltage at the 0 degree setting and qcw is the voltage at the 90 degree setting . the dc gain factor , gdc , can then be calculated . this gain factor can then be used as a correction factor to obtain calibrated waveforms of other signals . gdc is used for this purpose in step 10 . step 4 applies a modulated waveform as the microwave test signal 34 , generated , for example by means of the baseband time - domain transmitter 10 and communicated to the receiver 12 at input 38 in a test configuration . the baseband signal waveform is recorded by means of an analyzer ( dso or mta ) 44 , and the dc voltage is also recorded by means of voltmeter 56 . both the baseband waveform and the voltage are measured and recorded at two settings of the phase shifter 22 . the baseband waveform record should extend over more than one period of the periodic input microwave test signal 34 and 38 . the two phase shifter settings are 90 ° apart at the local oscillator frequency , and have the same two settings used in step 2 . idc is the voltage at the 0 ° setting and qdc is the voltage at the 90 ° setting , while i ( p ) is the baseband waveform at the 0 ° setting , and q ( p ) is the baseband waveform at the 90 ° setting . the time spacing between samples in the baseband waveforms i ( p ) and q ( p ) is defined as ts . step 5 resamples the baseband waveforms i ( p ) and q ( p ) using linear interpolation . other forms of interpolation could be applied such as using the sinc function , but the linear interpolation is adequate in most cases . the resampling should be done to achieve the desired number of samples , n , which should be an even number , and is preferentially a power of 2 , such as 1024 , for ease of subsequent discrete fourier transforms . the desired resampling interval , ts , is chosen such that nts = tp , where tp is one period of the periodic input signal . with these definitions , the formulas in step 4 are applied with inputs , i ( p ), q ( p ), ts , n , and ts . the outputs are ĩ ( n ) and { tilde over ( q )}( n ) and are the resampled baseband waveforms . step 6 subtracts the dc component from the resampled baseband waveforms . if all instruments were working perfectly , there would be no dc component to remove because the bias tee 54 physically separates the dc component from the rest of the baseband signal that is recorded by analyzer 44 . step 6 ensures that dc offsets in the baseband amplifier 42 and analyzer 44 are removed . the true dc component of the baseband signal will be added back into the baseband waveform in step 10 . the inputs for step 10 are the resampled waveforms ĩ ( n ) and { tilde over ( q )}( n ) and the outputs are the baseband waveforms ĩ a ( n ) and { tilde over ( q )} a ( n ) that are identical to the resampled waveforms ĩ ( n ) and { tilde over ( q )}( n ) except that the dc component has been subtracted out . step 7 extends the frequency range of the receiver transmission response to cover at least as great a frequency range as that covered by the discrete fourier transforms of the resampled waveforms . the lowpass equivalent receiver transmission response was obtained in step 1 . this measured receiver transmission response is defined as h meas ( ω meas ( m )), where the measurement extends over m frequencies from ω meas ( 0 ) to ω meas ( m − 1 ). the measured receiver transmission response is extended to ω =− π / ts in the negative frequency range , and to ω =+ π / ts in the positive frequency range to cover the nyquist frequencies of the resampled voltage waveforms . because there is no measured information about the receiver response outside the frequency range of ω meas ( 0 ) to ω meas ( m − 1 ), the extension is done by setting h ext ( ω ext ( l )) equal to a large value for frequencies outside the measured range . because the inverse of h ext ( ω ext ( l )) is used in step 7 calculations , making h ext ( ω ext ( l )) large outside the measured range of frequencies removes those frequencies from the final corrected waveforms . the inputs for this step 7 are the measured receiver response h meas ( ω meas ( m )), and the resampled waveform sampling time interval ts . the output is the receiver response extended in frequency h ext ( ω ext ( l )). step 8 resamples in the frequency domain , the extended receiver response h ext ( ω ext ( l )) to exactly match the discrete fourier transform of the baseband waveforms . a discrete fourier transform of a baseband waveform having n samples with a sampling interval ts , has frequency components given by ω ( k ) in step 8 . the equation in this step 8 is a linear interpolation of the receiver response h ext ( ω ext ( l )) to match the frequencies ω ( k ). the inputs to this step 8 are the waveform sampling interval , ts , the number of waveform samples n , and the extended receiver response h ext ( ω ext ( l )). the outputs are the discrete fourier transform frequencies ω ( k ), and the extended and resampled receiver response { tilde over ( h )}( ω ( k )). step 9 corrects the baseband waveforms ĩ a ( n ) and { tilde over ( q )} a ( n ) for the receiver response . the baseband waveforms are first transformed into the frequency domain using the discrete fourier transform . then the complex fourier coefficients are corrected for the receiver response by dividing by the extended and resampled receiver response . then the corrected frequency domain signal is transformed back into the time domain and the real and imaginary parts are extracted . the inputs for this step 9 are the resampled waveforms ĩ a ( n ) and { tilde over ( q )} a ( n ) and the extended and resampled receiver response { tilde over ( h )}( ω ( k )). the outputs of this step 9 are the real and imaginary parts of the corrected baseband waveforms ĩ c ( n ) and { tilde over ( q )} c ( n ). step 10 normalizes the corrected baseband waveforms in power as measured by means of a power meter . the measured signal power is defined as pmod . the rf normalization factor , grf is first calculated . the normalized waveform is then obtained using gdc from step 3 and grf . the inputs are pmod , gdc , idc , gdc , n , io , qo , ĩ a ( n ) and { tilde over ( q )} a ( n ). the outputs are the normalization factor grf , and the corrected and normalized real and imaginary waveforms ĩ p ( n ) and { tilde over ( q )} p ( n ). ĩ p ( n ) and { tilde over ( q )} p ( n ) are respectively the in - phase and quadrature components of the lpe waveform of the microwave test signal 34 . to measure the microwave response signal 38 , the dut must be inserted between the transmitter 10 and receiver 12 with the test signal 34 input to the dut and the dut output connected to the receiver 12 . then steps 4 through 10 need to be repeated . steps 1 through 3 do not need to be repeated because they are calibration steps , and the calibration will be good for a few hours . the above method is useful for characterization of arbitrary microwave signals 34 and nonlinear dut devices 36 . the method involves firstly characterizing the response of a downconverting receiver 12 comprising a downconverter using the prior ftd characterization method . secondly , the method is used to characterize an arbitrary test signal 34 , by removing the response of the receiver 12 from a sampled measurement . thirdly , the method is used to characterize a dut 36 by measuring the input and output lpe waveforms and removing the response of the downconverting receiver 12 from the lpe waveforms . the method can be applied to a wide range of arbitrary microwave signals 34 and test devices 36 . those skilled in the art can make enhancements , improvements , and modifications to the invention , and these enhancements , improvements , and modifications may nonetheless fall within the spirit and scope of the following claims .