the historical development of laser interferometers for application as gravitational - wave detectors   has involved the combination of relatively simple optical subsystems into more and more complex assemblies .
the individual elements that compose the interferometers , including mirrors , beam splitters , lasers , modulators , various polarising optics , photo detectors and so forth , are individually well described by relatively simple , mostly - classical physics .
complexity arises from the combination of multiple mirrors , beam splitters etc . into optical cavity systems ,
have narrow resonant features , and the consequent requirement to stabilise relative separations of the various components to sub - wavelength accuracy , and indeed in many cases to very small fractions of a wavelength .
thus , classical physics describes the interferometer techniques and the operation of current gravitational - wave detectors .
however , we note that at signal frequencies above a couple of hundreds of hertz , the sensitivity of current detectors is limited by the photon counting noise at the interferometer readout , also called shot - noise . the next generation systems such as advanced ligo
[ 23 , 5 ] , advanced virgo   and lcgt   are expected to operate in a regime where the quantum physics of both light and mirror motion couple to each other .
. the present review , in its first version , does not consider quantum effects but reserves them for future updates .
the components employed tend to behave in a linear fashion with respect to the optical field , i.e. , nonlinear optical effects need hardly be considered .
indeed , almost all aspects of the design of laser interferometers are dealt with in the linear regime .
therefore the underlying mathematics is relatively simple and many standard techniques are available , including those that naturally allow numerical solution by computer models .
such computer models are in fact necessary as the exact solutions can become quite complicated even for systems of a few components .
in practice , workers in the field rarely calculate the behaviour of the optical systems from first principles , but instead rely on various well - established numerical modelling techniques .
an example of software that enables modelling of either time - dependent or frequency - domain behaviour of interferometers and their component systems is finesse [ 22 , 19 ] .
this was developed by one of us ( af ) , has been validated in a wide range of situations , and was used to prepare the examples included in the present review .
the target readership we have in mind is the student or researcher who desires to get to grips with practical issues in the design of interferometers or component parts thereof .
for that reason , this review consists of sections covering the basic physics and approaches to simulation , intermixed with some practical examples . to make this as useful as possible
, the examples are intended to be realistic with sensible parameters reflecting typical application in gravitational wave detectors .
the examples , prepared using finesse , are designed to illustrate the methods typically applied in designing gravitational wave detectors .
we encourage the reader to obtain finesse and to follow the examples ( see appendix a ) .
as set out in very many works , gravitational - wave detectors strive to pick out signals carried by passing gravitational waves from a background of self - generated noise .
the principles of operation are set out at various points in the review , but in essence , the goal has been to prepare many photons , stored for as long as practical in the  arms of a laser interferometer ( traditionally the two arms are at right angles ) , so that tiny phase shifts induced by the gravitational waves form as large as possible a signal , when the light leaving the appropriate  port of the interferometer is detected and the resulting signal analysed . the evolution of gravitational - wave detectors can be seen by following their development from prototypes and early observing systems towards the advanced detectors , which are currently in the final stages of planning or early stages of construction .
starting from the simplest michelson interferometer , then by the application of techniques to increase the number of photons stored in the arms : delay lines , fabry - prot arm cavities [ 16 , 17 ] and power recycling .
the final step in the development of classical interferometry was the inclusion of signal recycling [ 41 , 30 ] , which , among other effects , allows the signal from a gravitational - wave signal of approximately - known spectrum to be enhanced above the noise .
reading out a signal from even the most basic interferometer requires minimising the coupling of local environmental effects to the detected output .
this is commonly achieved by suspending the mirrors etc . as pendulums , often multi - stage pendulums in series , and then applying closed - loop control to maintain the desired operating condition .
the careful engineering required to provide low - noise suspensions with the correct vibration isolation , and also low - noise actuation , is described in many works .
as the interferometer optics become more complicated , the resonance conditions , i.e. , the allowed combinations of inter - component path lengths required to allow the photon number in the interferometer arms to reach maximum , become more narrowly defined .
it is likewise necessary to maintain angular alignment of all components , such that beams required to interfere are correctly co - aligned .
typically the beams need to be aligned within a small fraction ( and sometimes a very small fraction ) of the far - field diffraction angle , and the requirement can be in the low nanoradian range for km - scale detectors [ 44 , 21 ] . therefore , for each optical component there is typically one longitudinal ( i.e. , along the direction of light propagation ) , plus two angular degrees of freedom ( pitch and yaw about the longitudinal axis ) .
a complex interferometer can consist of up to around seven highly sensitive components and so there can be of order 20 degrees of freedom to be measured and controlled [ 3 , 57 ] .
although the light fields are linear , the coupling between the position of a mirror and the complex amplitude of the detected light field typically shows strongly nonlinear dependence on mirror positions due to the sharp resonance features exhibited by cavity systems .
however , the fields do vary linearly or at least smoothly close to the desired operating point .
so , while well - understood linear control theory suffices to design the control system needed to maintain the optical configuration at its operating point , bringing the system to that operating condition is often a separate and more challenging nonlinear problem . in the current version of this work
control systems require actuators , and those employed are typically electrical - force transducers that act on the suspended optical components , either directly or  to provide enhanced noise rejection  at upper stages of multi - stage suspensions .
the transducers are normally coil - magnet actuators , with the magnets on the moving part , or , less frequently , electrostatic actuators of varying design .
the actuators are frequently regarded as part of the mirror suspension subsystem and are not discussed in the current work . to give order to our review we consider the main physics describing the operation of the basic optical components ( mirrors , beam splitters , modulators , etc . ) required to construct interferometers . although all of the relevant physics is generally well known and not new , we take it as a starting point that permits the introduction of notation and conventions .
it is also true that the interferometry employed for gravitational - wave detection has a different emphasis than other interferometer applications . as a consequence , descriptions or examples of a number of crucial optical properties for gravitational wave detectors
the purpose of this first version of the review is especially to provide a coherent theoretical framework for describing such effects .
with the basics established , it can be seen that the interferometer configurations that have been employed in gravitational - wave detection may be built up and simulated in a relatively straightforward manner . as mentioned above
, we do not address the newer physics associated with operation at or beyond the standard quantum limit .
the standard quantum limit [ 10 , 32]squeezing [ 38 , 53]quantum nondemolition interferometry [ 9 , 24 ] 
 these matters are to be included in a future revision of this review . the standard quantum limit [ 10 , 32 ] quantum nondemolition interferometry [ 9 , 24 ] the main optical systems of interferometric gravitational - wave detectors
the stability is achieved through a mixture of passive isolation systems and active feedback control . in particular , the light sources are some of the most stable , low - noise continuous - wave laser systems so that electromagnetic fields can be assumed to be essentially monochromatic .
additional frequency components can be modelled as small modulations ( in amplitude or phase ) .
the laser beams are well collimated , propagate along a well - defined optical axis and remain always very much smaller than the optical elements they interact with .
therefore , these beams can be described as paraxial and the well - known paraxial approximations can be applied .
it is useful to first derive a mathematical model based on monochromatic , scalar , plane waves . as it turns out ,
a more detailed model including the polarisation and the shape of the laser beam as well as multiple frequency components , can be derived as an extension to the plane - wave model . a plane electromagnetic wave
is typically described by its electric field component : with e0 as the ( constant ) field amplitude in v / m , \documentclass[12pt]{minimal } 
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				 \begin{document}${\vec e_p}$\end{document } the unit vector in the direction of polarisation , such as , for example , \documentclass[12pt]{minimal } 
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				 \begin{document}${\mathscr i}$\end{document}-polarised light ,  the angular oscillation frequency of the wave , and \documentclass[12pt]{minimal } 
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				 \begin{document}$\vec k = { \vec e_k}\omega / c$\end{document } the wave vector pointing the in the direction of propagation .
the absolute phase  only becomes meaningful when the field is superposed with other light fields . 

figure 1 in this document we will consider waves propagating along the optical axis given by the z - axis , so that \documentclass[12pt]{minimal } 
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				 \begin{document}$\vec k\vec r = kz$\end{document}. for the moment we will ignore the polarisation and use scalar waves , which can be written as 1\documentclass[12pt]{minimal } 
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				 \begin{document}$$e(z , t ) = { e_0}\cos ( \omega t - kz + \varphi){.}$$\end{document } further , in this document we use complex notation , i.e. , 2\documentclass[12pt]{minimal } 
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				 \begin{document}$$e = \re\{e^{\prime}\ } \quad { \rm{with}}\quad e^{\prime } = e_0^{\prime}\exp ( { \rm{i}}(\omega t - kz)).$$\end{document } this has the advantage that the scalar amplitude and the phase  can be given by one , now complex , amplitude e0 = e0 exp(i ) we will use this notation with complex numbers throughout . for clarity we will simply use the unprimed letters for the auxiliary field . in particular , we will use the letter e and also a and b to denote complex electric - field amplitudes . but remember that , for example , in e = e0 exp(i kz ) neither e nor e0 are physical quantities . only the real part of e exists and deserves the name field amplitude . in most cases we are
either interested in the fields at one particular location , for example , on the surface of an optical element , or we want to know the fields at all places in the interferometer but at one particular point in time .
the latter is usually true for the steady state approach : assuming that the interferometer is in a steady state , all solutions must be independent of time so that we can perform all computations at t = 0 without loss of generality . in that case , the scalar plane wave can be written as 3\documentclass[12pt]{minimal } 
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				 \begin{document}$$e = { e_0}\exp ( -{\rm{i}}\;kz).$$\end{document } the frequency domain is of special interest as numerical models of gravitational - wave detectors tend to be much faster to compute in the frequency domain than in the time domain .
when an electromagnetic wave interacts with an optical system , all of its parameters can be changed as a result . typically optical components are designed such that , ideally , they only affect one of the parameters , i.e. , either the amplitude or the polarisation or the shape .
typically , the optical components are described in the simplest possible way , as illustrated by the use of abstract schematics such as those shown in figure 2 . 

figure 2this set of figures introduces an abstract form of illustration , which will be used in this document .
the top figure shows a typical example taken from the analysis of an optical system : an incident field ein is refiected and transmitted by a semi - transparent mirror ; there might be the possibility of second incident field ein2 .
the lower left figure shows the abstract form we choose to represent the same system .
the lower right figure depicts how this can be extended to include a beam splitter object , which connects two optical axes .
this set of figures introduces an abstract form of illustration , which will be used in this document .
the top figure shows a typical example taken from the analysis of an optical system : an incident field ein is refiected and transmitted by a semi - transparent mirror ; there might be the possibility of second incident field ein2 .
the lower left figure shows the abstract form we choose to represent the same system .
the lower right figure depicts how this can be extended to include a beam splitter object , which connects two optical axes .
the core optical systems of current interferometric gravitational interferometers are composed of two building blocks : a ) resonant optical cavities , such as fabry - prot resonators , and b ) beam splitters , as in a michelson interferometer .
in other words , the laser beam is either propagated through a vacuum system or interacts with a partially - reflecting optical surface .
the term optical surface generally refers to a boundary between two media with possibly different indices of refraction n , for example , the boundary between air and glass or between two types of glass .
a real fused silica mirror in an interferometer features two surfaces , which interact with a reffected or transmitted laser beam .
however , in some cases , one of these surfaces has been treated with an anti - reffection ( ar ) coating to minimise the effect on the transmitted beam . the terms mirror and beam splitter
we define real amplitude coefficients for reflection and transmission r and t , with 0  r , t  1 , so that the field amplitudes can be written as the /2 phase shift upon transmission ( here given by the factor i ) refers to a phase convention explained in section 2.4 .
the free propagation of a distance d through a medium with index of refraction n can be described with the following set of equations : in the following we use n = 1 for simplicity .
note that we use above relations to demonstrate various mathematical methods for the analysis of optical systems .
however , refined versions of the coupling equations for optical components , including those for spaces and mirrors , are also required , see , for example .
the linear optical resonator , also called a cavity is formed by two partially - transparent mirrors , arranged in parallel as shown in figure 5 .
this simple setup makes a very good example with which to illustrate how a mathematical model of an interferometer can be derived , using the equations introduced in section 2.1 .
the cavity is defined by a propagation length d ( in vacuum ) , the amplitude reflectivities r1 , r2 and the amplitude transmittances t1 , t2 . the amplitude at each point in the cavity can be computed simply as the superposition of fields .
the entire set of equations can be written as 4\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { a_1 } = { \rm{i}}\;{t_1}{a_0 } + { r_1}a_3^\prime \quad}\\ { a_1^\prime = \exp ( - { \rm{i}}\;kd){a_1}}\\ { { a_2 } = { \rm{i}}\;{t_2}a_1^\prime \quad \quad \quad}\\ { { a_3 } = { r_2}a_1^\prime \quad \quad \quad \quad}\\ { a_3^\prime = \exp ( - { \rm{i}}\;kd){a_3}}\\ { { a_4 } = { r_1}{a_0 } + { \rm{i}}\;{t_1}a_3^\prime \quad}\\\end{array}$$\end{document } the circulating field impinging on the first mirror ( surface ) a3 can now be computed as 5\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { a_3^\prime = \exp ( - { \rm{i}}\,kd){a_3 } = \exp ( - { \rm{i}}\,kd){r_2}a_1^\prime = \exp ( - { \rm{i}}\,2kd){r_2}{a_1 } } \\ { = \exp ( - { \rm{i}}\,{\rm{2}}kd)\,{r_2}({\rm{i}}\,{t_1}{a_0 } + { r_1}a_3^\prime).\quad \quad \quad \quad \quad \quad \quad \,\,\ , } \\ \end{array}$$\end{document } this then yields 6\documentclass[12pt]{minimal } 
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				 \begin{document}$$a_3^{\prime } = { a_0}{{{\rm{i}}\;{r_2}{t_1}\exp ( - { \rm{i\;2}}kd ) } \over { 1 - { r_1}{r_2}\exp ( - { \rm{i\;2}}kd)}}.$$\end{document } we can directly compute the reflected field to be 7\documentclass[12pt]{minimal } 
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				 \begin{document}$${a_4 } = { a_0}\left({{r_1 } - { { { r_2}t_1 ^ 2\exp ( - { \rm{i\;2}}kd ) } \over { 1 - { r_1}{r_2}\exp ( - { \rm{i\;2}}kd ) } } } \right ) = { a_0}\left({{{{r_1 } - { r_2}(r_1 ^ 2 + t_1 ^ 2)\exp ( - { \rm{i\;2}}kd ) } \over { 1 - { r_1}{r_2}\exp ( - { \rm{i\;2}}kd ) } } } \right),$$\end{document } while the transmitted field becomes 8\documentclass[12pt]{minimal } 
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				 \begin{document}$${a_2 } = { a_0}{{- { t_1}{t_2}\exp ( - { \rm{i}}\;kd ) } \over { 1 - { r_1}{r_2}\exp ( - { \rm{i\;2}}kd)}}.$$\end{document } the properties of two mirror cavities will be discussed in more detail in section 5.1 .
computations that involve sets of linear equations as shown in section 2.2 can often be done or written efficiently with matrices .
two methods of applying matrices to coupling field amplitudes are demonstrated below , using again the example of a two mirror cavity .
the mirror coupling as given in figure 3 becomes and the amplitude coupling at a 
space , as given in figure 4 , can be written as in these examples the matrix simply transforms the 
figure 3 
 figure 4 an obvious application of the matrices introduced above would be to construct a large matrix for an extended optical system appropriate for computerisation .
the set of linear equations for a mirror would expand to 9\documentclass[12pt]{minimal } 
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				 \begin{document}$$\left({\begin{array}{*{20}c } 1 & 0 & 0 & 0 \\ { - { \rm{i}}\;t } & 1 & 0 & { - r } \\ 0 & 0 & 1 & 0 \\ { - r } & 0 & { - { \rm{i}}\;t } & 1 \\ \end{array } } \right)\left({\begin{array}{*{20}c } { { a_1 } } \\ { { a_2 } } \\ { { a_3 } } \\ { { a_4 } } \\
\end{array } } \right ) = \left({\begin{array}{*{20}c } { { a_1 } } \\ 0 \\ { { a_3 } } \\ 0 \\ \end{array } } \right ) = { m_{{\rm{system}}}}{\vec a_{{\rm{sol } } } } = { \vec a_{{\rm{input}}}},$$\end{document } where the input vector1 
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				 \begin{document}${{\vec a}_{{\rm{input}}}}$\end{document } has non - zero values for the impinging fields and \documentclass[12pt]{minimal } 
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				 \begin{document}${{\vec a}_{{\rm{sol}}}}$\end{document } is the 
solution vector , i.e. , after solving the system of equations the amplitudes of the impinging as well as those of the outgoing fields are stored in that vector . as an example we apply this method to the two mirror cavity .
the system matrix for the optical setup shown in figure 5 becomes 10\documentclass[12pt]{minimal } 
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				 \begin{document}$$\left({\begin{array}{*{20}c } 1 & 0 & 0 & 0 & 0 & 0 & 0 \\ { - { \rm{i}}\;{t_1 } } & 1 & 0 & { - { r_1 } } & 0 & 0 & 0 \\ { - { r_1 } } & 0 & 1 & { - { \rm{i}}\;{t_1 } } & 0 & 0 & 0 \\ 0 & 0 & 0 & 1 & 0 & 0 & { - { e^{- { \rm{i}}\;kd } } } \\ 0 & { - { e^{- { \rm{i}}\;kd } } } & 0 & 0 & 1 & 0 & 0 \\ 0 & 0 & 0 & 0 & { - { \rm{i}}\;{t_1 } } & 1 & 0 \\ 0 & 0 & 0 & 0 & 0 & { - { r_2 } } & 1 \\ \end{array } } \right)\left({\begin{array}{*{20}c } { { a_0 } } \\ { { a_1 } } \\ { { a_4 } } \\ { a_3^\prime } \\
{ a_1^\prime } \\ { { a_2 } } \\ { { a_3 } } \\
\end{array } } \right ) = \left({\begin{array}{*{20}c } { { a_0 } } \\ 0 \\ 0 \\ 0 \\ 0 \\ 0 \\ 0 \\ \end{array } } \right)$$\end{document } this is a sparse matrix .
sparse matrices are an important subclass of linear algebra problems and many efficient numerical algorithms for solving sparse matrices are freely available ( see , for example , ) .
the advantage of this method of constructing a single matrix for an entire optical system is the direct access to all field amplitudes .
it also stores each coupling coefficient in one or more dedicated matrix elements , so that numerical values for each parameter can be read out or changed after the matrix has been constructed and , for example , stored in computer memory .
the obvious disadvantage is that the size of the matrix quickly grows with the number of optical elements ( and with the degrees of freedom of the system , see , for example , section 7 ) . 

further , the resulting cavity is characterised by its length d. light field amplitudes are shown and identified by a variable name , where necessary to permit their mutual coupling to be computed .
further , the resulting cavity is characterised by its length d. light field amplitudes are shown and identified by a variable name , where necessary to permit their mutual coupling to be computed .
the following method is probably most useful for analytic computations , or for optimisation aspects of a numerical computation .
the idea behind the scheme , which is used for computing the characteristics of dielectric coatings [ 28 , 40 ] and has been demonstrated for analysing gravitational wave detectors , is to rearrange equations as in figure 6 and figure 7 such that the overall matrix describing a series of components can be obtained by multiplication of the component matrices . in order to achieve this
, the coupling equations have to be re - ordered so that the input vector consists of two field amplitudes at one side of the component . for the mirror ,
this gives a coupling matrix of 11\documentclass[12pt]{minimal } 
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				 \begin{document}$$\left({\begin{array}{*{20}c } { { a_1}}\\ { { a_4}}\\ \end{array } } \right ) = { { \rm{i } } \over t}\left({\begin{array}{*{20}c } { - 1 } & \quad r\\ { - r } & { { r^2 } + { t^2}}\\ \end{array } } \right)\left({\begin{array}{*{20}c } { { a_2}}\\ { { a_3}}\\ \end{array } } \right).$$\end{document } in the special case of the lossless mirror this matrix simplifies as we have r + t = r + t = 1 .
the space component would be described by the following matrix : 12\documentclass[12pt]{minimal } 
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				 \begin{document}$$\left({\begin{array}{*{20}c } { { a_1 } } \\ { { a_4 } } \\
\end{array } } \right ) = \left({\begin{array}{*{20}c } { \exp ( { \rm{i}}\;kd ) } & { \quad 0 } \\ { \quad 0 } & { \exp ( - { \rm{i}}\;kd ) } \\ \end{array } } \right)\left({\begin{array}{*{20}c } { { a_2 } } \\ { { a_3 } } \\\end{array } } \right).$$\end{document } with these matrices we can very easily compute a matrix for the cavity with two lossless mirrors as 13\documentclass[12pt]{minimal } 
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				 \begin{document}$${m_{{\rm{cav } } } } = { m_{{\rm{mirror}}1 } } \times { m_{{\rm{space } } } } \times { m_{{\rm{mirror}}2}}$$\end{document } 
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				 \begin{document}$$= { { - 1 } \over { { t_1}{t_2}}}\left({\begin{array}{*{20}c } { { e^ + } - { r_1}{r_2}{e^ - } } & { \quad - { r_2}{e^ + } + { r_1}{e^ -}}\\ { - { r_2}{e^ - } + { r_1}{e^ + } } & { { e^ - } - { r_1}{r_2}{e^ + } } \\
\end{array } } \right),$$\end{document } with e = exp(i kd ) and e = exp(ikd ) .
the system of equation describing a cavity shown in equation ( 4 ) can now be written more compactly as 15\documentclass[12pt]{minimal } 
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				 \begin{document}$$\left({\begin{array}{*{20}c } { { a_0}}\\ { { a_4}}\\ \end{array } } \right ) = { { - 1 } \over { { t_1}{t_2}}}\left({\begin{array}{*{20}c } { { e^ + } - { r_1}{r_2}{e^ - } } & { \quad - { r_2}{e^ + } + { r_1}{e^ -}}\\ { - { r_2}{e^ - } + { r_1}{e^ + } } & { { e^ - } - { r_1}{r_2}{e^ + } } \\ \end{array } } \right)\left({\begin{array}{*{20}c } { { a_2}}\\ 0\\ \end{array } } \right).$$\end{document } this allows direct computation of the amplitude of the transmitted field resulting in 16\documentclass[12pt]{minimal } 
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				 \begin{document}$${a_2 } = { a_0}{{- { t_1}{t_2}\exp ( - { \rm{i}}\;kd ) } \over { 1 - { r_1}{r_2}\exp ( - { \rm{i\;2}}kd)}},$$\end{document } which is the same as equation ( 8) . 
 figure 6 
 figure 7 the advantage of this matrix method is that it allows compact storage of any series of mirrors and propagations , and potentially other optical elements , in a single 2  2 matrix .
the disadvantage inherent in this scheme is the lack of information about the field amplitudes inside the group of optical elements .
the magnitude and phase of reflection at a single optical surface can be derived from maxwell s equations and the electromagnetic boundary conditions at the surface , and in particular the condition that the field amplitudes tangential to the optical surface must be continuous .
thus , for a field impinging on an optical surface under normal incidence we can give the reflection coefficient as 17\documentclass[12pt]{minimal } 
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				 \begin{document}$$r = { { { n_1 } - { n_2 } } \over { { n_1 } + { n_2}}},$$\end{document } with n1 and n2 the indices of refraction of the first and second medium , respectively .
the transmission coefficient for a lossless surface can be computed as t = 1  r. we note that the phase change upon reflection is either 0 or 180 , depending on whether the second medium is optically thinner or thicker than the first .
it is not shown here but fresnel s equations can also be used to show that the phase change for the transmitted light at a lossless surface is zero .
this contrasts with the definitions given in section 2.1 ( see figure ( 3)ff . ) , where the phase shift upon any reflection is defined as zero and the transmitted light experiences a phase shift of /2 .
the following section explains the motivation for the latter definition having been adopted as the common notation for the analysis of modern optical systems .
modern mirrors and beam splitters that make use of dielectric coatings are complex optical systems , see figure 8 whose reflectivity and transmission depend on the multiple interference inside the coating layers and thus on microscopic parameters .
the phase change upon transmission or reflection depends on the details of the applied coating and is typically not known . in any case , the knowledge of an absolute value of a phase change is typically not of interest in laser interferometers because the absolute positions of the optical components are not known to sub - wavelength precision .
instead the relative phase between the incoming and outgoing beams is of importance . in the following
we demonstrate how constraints on these relative phases , i.e. , the phase relation between the beams , can be derived from the fundamental principle of power conservation . to do this we consider a michelson interferometer , as shown in figure 9 , with perfectly - reflecting mirrors .
we can say that the phase change upon transmission should be the same in both directions .
however , the phase change on reflection might be different for either direction , thus , we write for the reflection at the front and for the reflection at the back of the beam splitter . 

figure 8this sketch shows a mirror or beam splitter component with dielectric coatings and the photograph shows some typical commercially available examples .
most mirrors and beam splitters used in optical experiments are of this type : a substrate made from glass , quartz or fused silica is coated on both sides .
the reflective coating defines the overall reflectivity of the component ( anything between r  1 and r  0 , while the anti - reflective coating is used to reduce the reflection at the second optical surface as much as possible so that this surface does not influence the light .
please note that the drawing is not to scale , the coatings are typically only a few microns thick on a several millimetre to centimetre thick substrate . 

figure 9the relation between the phase of the light field amplitudes at a beam splitter can be computed assuming a michelson interferometer , with arbitrary arm length but perfectly - reflecting mirrors .
the incoming field e0 is split into two fields e1 and e2 which are reflected atthe end mirrors and return to the beam splitter , as e3 and e4 , to be recombined into two outgoing fields .
these outgoing fields e5 and e6 are depicted by two arrows to highlight that these are the sum of the transmitted and reflected components of the returning fields .
we can derive constraints for the phase of e1 and e2 with respect to the input field e0 from the conservation of energy : |e0| = |e5| + |e6| .
this sketch shows a mirror or beam splitter component with dielectric coatings and the photograph shows some typical commercially available examples .
most mirrors and beam splitters used in optical experiments are of this type : a substrate made from glass , quartz or fused silica is coated on both sides . the reflective coating defines the overall reflectivity of the component ( anything between r  1 and r  0 , while the anti - reflective coating is used to reduce the reflection at the second optical surface as much as possible so that this surface does not influence the light .
please note that the drawing is not to scale , the coatings are typically only a few microns thick on a several millimetre to centimetre thick substrate .
the relation between the phase of the light field amplitudes at a beam splitter can be computed assuming a michelson interferometer , with arbitrary arm length but perfectly - reflecting mirrors .
the incoming field e0 is split into two fields e1 and e2 which are reflected atthe end mirrors and return to the beam splitter , as e3 and e4 , to be recombined into two outgoing fields .
these outgoing fields e5 and e6 are depicted by two arrows to highlight that these are the sum of the transmitted and reflected components of the returning fields .
we can derive constraints for the phase of e1 and e2 with respect to the input field e0 from the conservation of energy : |e0| = |e5| + |e6| .
then the electric fields can be computed as 18\documentclass[12pt]{minimal } 
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				 \begin{document}$${e_1 } = r\;{e_0}\;{e^{{\rm{i}}{\varphi _ { r1}}}};\quad { e_2 } = t\;{e_0}\;{e^{{\rm{i}}{\varphi _ t}}}.$$\end{document } we do not know the length of the interferometer arms .
thus , we introduce two further unknown phases : 1 for the total phase accumulated by the field in the vertical arm and 2 for the total phase accumulated in the horizontal arm .
the fields impinging on the beam splitter compute as 19\documentclass[12pt]{minimal } 
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				 \begin{document}$${e_3 } = r\;{e_0}\;{e^{{\rm{i}}({\varphi _ { r1 } } + { \phi _ 1})}};\quad { e_4 } = t\;{e_0}\;{e^{{\rm{i}}({\varphi _ t } + { \phi _ 2})}}.$$\end{document } the outgoing fields are computed as the sums of the reflected and transmitted components : 20\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { e_5 } = \ { e_0}(r\ { e^{{\rm{i}}(2{\varphi _ { r1 } } + { \phi _ 1 } ) } } + t\ { e^{{\rm{i}}(2{\varphi _ t } + { \phi _ 2})}}).}\quad\\ { { e_6 } = \ { e_0}\;rt(\ { e^{{\rm{i}}({\varphi _ t } + { \varphi _ { r1 } } + { \phi _ 1 } ) } } + { e^{{\rm{i}}({\varphi _ t } + { \varphi _ { r2 } } + { \phi _ 2})}}),}\\ \end{array}$$\end{document } with r = r and t = t. it will be convenient to separate the phase factors into common and differential ones .
we can write 21\documentclass[12pt]{minimal } 
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				 \begin{document}$${e_5 } = { e_0}\;{e^{{\rm{i}}{\alpha _ + } } } ( r{e^{{\rm{i}}{\alpha _ - } } } + t{e^{{\rm{- i}}{\alpha _ -}}}),$$\end{document } with 22\documentclass[12pt]{minimal } 
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				 \begin{document}$${\alpha _ + } = { \varphi _ { r1 } } + { \varphi _ t } + { 1 \over 2}({\phi _ 1 } + { \phi _
2});\quad { \alpha _ - } = { \varphi _ { r1 } } - { \varphi _ t } + { 1 \over 2}({\phi _ 1 } - { \phi _ 2}),$$\end{document } and similarly 23\documentclass[12pt]{minimal } 
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				 \begin{document}$${e_6 } = { e_0}\;rt\;{e^{{\rm{i}}{\beta _ { + } } } } 2\cos ( { \beta _ -}),$$\end{document } with 24\documentclass[12pt]{minimal } 
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				 \begin{document}$${\beta _ + } = { \varphi _ t } + { 1 \over 2}({\varphi _ { r1 } } + { \varphi _ { r2 } } + { \phi _ 1 } + { \phi _ 2});\quad { \beta _ - } = { 1 \over 2}({\varphi _ { r1 } } - { \varphi _ { r2 } }
+ { \phi _ 1 } - { \phi _ 2}){.}$$\end{document } for simplicity we now limit the discussion to a 50:50 beam splitter with \documentclass[12pt]{minimal } 
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				 \begin{document}$r = t = 1/\sqrt 2$\end{document } , for which we can simplify the field expressions even further : 25\documentclass[12pt]{minimal } 
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				 \begin{document}$${e_5 } = { e_0}\ ; { e^{{\rm{i}}{\alpha _ + } } } \cos ( { \alpha _ -});\quad { e_6 } = { e_0 } \;{e^{{\rm{i}}{\beta _ + } } } \cos ( { \beta _ -}){.}$$\end{document } conservation of energy requires that |e0| = |e5| + |e6| , which in turn requires 26\documentclass[12pt]{minimal } 
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				 \begin{document}$${\cos ^2}({\alpha _ - } ) + { \cos ^2}({\beta _ - } ) = 1,$$\end{document } which is only true if 27\documentclass[12pt]{minimal } 
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				 \begin{document}$${\alpha _ - } - { \beta _ - } = ( 2n + 1){\pi \over 2},$$\end{document } with n as in integer ( positive , negative or zero ) .
this gives the following constraint on the phase factors 28\documentclass[12pt]{minimal } 
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				 \begin{document}$${1 \over 2}({\varphi _ { r1 } } + { \varphi _ { r2 } } ) - { \varphi _ t } = ( 2n + 1){\pi \over 2}.$$\end{document } one can show that exactly the same condition results in the case of arbitrary ( lossless ) reflectivity of the beam splitter .
we can test whether two known examples fulfill this condition . if the beam - splitting surface is the front of a glass plate we know that t = 0 , r1 =  r2 = 0 , which conforms with equation ( 28 ) .
if we consider the cavity as an optical  black box , it also splits any incoming beam into a reflected and transmitted component , like a mirror or beam splitter .
further we know that a symmetric resonator must give the same results for fields injected from the left or from the right .
thus , the phase factors upon reflection must be equal r = r1 = r2 .
the reflection and transmission coefficients are given by equations ( 7 ) and ( 8) as 29\documentclass[12pt]{minimal } 
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				 \begin{document}$${r_{{\rm{cav } } } } = \left({{r_1 } - { { { r_2}t_1 ^ 2\exp ( - { \rm{i}}\;{\rm{2}}kd ) } \over { 1 - { r_1}{r_2}\exp ( - { \rm{i}}\;{\rm{2}}kd ) } } } \right),$$\end{document } and 30\documentclass[12pt]{minimal } 
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				 \begin{document}$${t_{{\rm{cav } } } } = { { - { t_1}{t_2}\exp ( - { \rm{i}}\;kd ) } \over { 1 - { r_1}{r_2}\exp ( - { \rm{i\;2}}kd)}}.$$\end{document } we demonstrate a simple case by putting the cavity on resonance ( kd = n ) .
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				 \begin{document}$${r_{{\rm{cav } } } } = \left({{r_1 } - { { { r_2}t_1 ^ 2 } \over { 1 - { r_1}{r_2 } } } } \right);\quad { t_{{\rm{cav } } } } = { { { \rm{i}}\;{t_1}{t_2 } } \over { 1 - { r_1}{r_2}}},$$\end{document } with rcav being purely real and tcav imaginary and thus t = /2 and r = 0 which also agrees with equation ( 28 ) . in most cases we neither know nor care about the exact phase factors .
instead we can pick any set which fulfills equation ( 28 ) . for this document
we have chosen to use phase factors equal to those of the cavity , i.e. , t = /2 and r = 0 , which is why we write the reflection and transmission at a mirror or beam splitter as 32\documentclass[12pt]{minimal } 
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				 \begin{document}$${e_{{\rm{refl } } } } = r\;{e_{0\quad}}{\rm{and}}\quad { e_{{\rm{trans } } } } = { \rm{i}}\;t\;{e_0}.$$\end{document } in this definition r and t are positive real numbers satisfying r + t = 1 for the lossless case .
please note that we only have the freedom to chose convenient phase factors when we do not know or do not care about the details of the optical system , which performs the beam splitting .
if instead the details are important , for example when computing the properties of a thin coating layer , such as anti - reflex coatings , the proper phase factors for the respective interfaces must be computed and used .
the resonance condition inside an optical cavity and the operating point of an interferometer depends on the optical path lengths modulo the laser wavelength , i.e. , for light from an nd : yag laser length differences of less than 1 m are of interest , not the full magnitude of the distances between optics . on the other hand ,
several parameters describing the general properties of an optical system , like the finesse or free spectral range of a cavity ( see section 5.1 ) depend on the macroscopic distance and do not change significantly when the distance is changed on the order of a wavelength .
this illustrates that the distance between optical components might not be the best parameter to use for the analysis of optical systems .
furthermore , it turns out that in numerical algorithms the distance may suffer from rounding errors .
the cavity length is approximately 3 km , the wavelength is on the order of 1 m , the mirror positions are actively controlled with a precision of 1 m and the detector sensitivity can be as good as 10 m , measured on  10 ms timescales ( i.e. , many samples of the data acquisition rate ) . the floating point accuracy of common , fast numerical algorithms is typically not better than 10 .
if we were to store the distance between the cavity mirrors as such a floating point number , the accuracy would be limited to 3 pm , which does not even cover the accuracy of the control systems , let alone the sensitivity . 

figure 10illustration of an arm cavity of the virgo gravitational - wave detector : the macroscopic length l of the cavity is approximately 3 km , while the wavelength of the nd : yag laser is   1 m . the resonance condition is only affected by the microscopic position of the wave nodes with respect to the mirror surfaces and not by the macroscopic length , i.e. , displacement of one mirror by x = /2 re - creates exactly the same condition . however , other parameters of the cavity , such as the finesse , only depend on the macroscopic length l and not on the microscopic tuning .
illustration of an arm cavity of the virgo gravitational - wave detector : the macroscopic length l of the cavity is approximately 3 km , while the wavelength of the nd : yag laser is   1 m . the resonance condition is only affected by the microscopic position of the wave nodes with respect to the mirror surfaces and not by the macroscopic length , i.e. , displacement of one mirror by x = /2 re - creates exactly the same condition . however , other parameters of the cavity , such as the finesse , only depend on the macroscopic length l and not on the microscopic tuning .
a simple and elegant solution to this problem is to split a distance d between two optical components into two parameters : one is the macroscopic 
length l , defined as the multiple of a constant wavelength 0 yielding the smallest difference to d. the second parameter is the microscopic tuning t that is defined as the remaining difference between l and d , i.e. , d = l + t. typically , 0 can be understood as the wavelength of the laser in vacuum , however , if the laser frequency changes during the experiment or multiple light fields with different frequencies are used simultaneously , a default constant wavelength must be chosen arbitrarily .
please note that usually the term  in any equation refers to the actual wavelength at the respective location as  = 0/n with n the index of refraction at the local medium .
we have seen in section 2.1 that distances appear in the expressions for electromagnetic waves in connection with the wave number , for example , 33\documentclass[12pt]{minimal } 
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				 \begin{document}$${e_{\rm{2 } } } = { e_{1\quad}}\exp ( - { \rm{i}}\;kz)\;.$$\end{document } thus , the difference in phase between the field at z = z1 and z = z1 + d is given as 34\documentclass[12pt]{minimal } 
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				 \begin{document}$$\varphi = - kd.$$\end{document } we recall that k = 2/ = /c .
we can define 0 = 2 c/0 and k0 = 0/c . for any given wavelength  we can write the corresponding frequency as a sum of the default frequency and a difference frequency  = 0 + . using these definitions , we can rewrite equation ( 34 ) with length and tuning as 35\documentclass[12pt]{minimal } 
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				 \begin{document}$$- \varphi = kd = { { { \omega _ 0}l } \over c } + { { \delta \omega l } \over c } + { { { \omega _ 0}t } \over c } + { { \delta \omega t } \over c}.$$\end{document } the first term of the sum is always a multiple of 2 , which is equivalent to zero .
the last term of the sum is the smallest , approximately of the order   10 . for typical values of l  1 m , t < 1 m and  < 2  100 mhz we find that 36\documentclass[12pt]{minimal } 
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				 \begin{document}$${{{\omega _ 0}l } \over c } = 0,\quad { { \delta \omega l } \over c}\underset{\approx } { < } 2,\quad { { { \omega _ 0}t } \over c } \underset{\approx } { < } 6,\quad { { \delta \omega t } \over c } \underset{\approx } { < } 2\;\;\;{10^{- 6}},$$\end{document } which shows that the last term can often be ignored . we can also write the tuning directly as a phase .
we define as the dimensionless tuning 37\documentclass[12pt]{minimal } 
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				 \begin{document}$$\phi = { \omega _ 0}t / c.$$\end{document } this yields 38\documentclass[12pt]{minimal } 
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				 \begin{document}$$\exp \left({{\rm{i}}{\omega \over c}t }
\right ) = \exp \left({{\rm{i}}{{{\omega _ 0 } } \over c}t{\omega \over { { \omega _ 0 } } } } \right ) = \exp \left({{\rm{i}}{\omega \over { { \omega _ 0}}}\phi } \right).$$\end{document } the tuning  is given in radian with 2 referring to a microscopic distance of one wavelength2 0 .
finally , we can write the following expression for the phase difference between the light field taken at the end points of a distance d : 39\documentclass[12pt]{minimal } 
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				 \begin{document}$$\varphi = - kd = - \left({{{\delta \omega l } \over c } + \phi { \omega \over { { \omega _ 0 } } } } \right),$$\end{document } or if we neglect the last term from equation ( 36 ) we can approximate ( /0  1 ) to obtain 40\documentclass[12pt]{minimal } 
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				 \begin{document}$$\varphi \approx - \left({{{\delta \omega l } \over c } + \phi } \right).$$\end{document } this convention provides two parameters l and  that can describe distances with a markedly improved numerical accuracy . in addition , this definition often allows simplification of the algebraic notation of interferometer signals . by convention
we associate a length l with the propagation through free space , whereas the tuning will be treated as a parameter of the optical components .
if , for example , a cavity is to be resonant to the laser light , the tunings of the mirrors have to be the same whereas the length of the space in between can be arbitrary . using the definitions for length and tunings we can rewrite the coupling equations for mirrors and spaces introduced in section 2.1 as follows .
the mirror coupling becomes ( compare this to figure 6 ) , and the amplitude coupling for a 
space , formally written as in figure 7 , is now written as 
 figure 11 
 figure 12 we use finesse to plot the amplitudes of the light fields transmitted and reflected by a mirror ( given by a single surface ) .
initially , the mirror has a power reflectance and transmittance of r = t = 0.5 and is , thus , lossless . for the plot in figure 13 we tune the transmittance from 0.5 to 0 .
since we do not explicitly change the reflectivity , r remains at 0.5 and the mirror loss increases instead , which is shown by the trace labelled 
the plot also shows the phase convention of a 90 phase shift for the transmitted light . 

finesse input file for  mirror reflectivity and transmittance 
  this finesse file demonstrates the conventions for lengths and microscopic positions introduced in section 2.5 .
the top trace in figure 14 depicts the phase change of a beam reflected by a beam splitter as the function of the beam splitter tuning . by changing the tuning from 0 to 180
the beam splitter is moved forward and shortens the path length by one wavelength , which by convention increases the light phase by 360. on the other hand , if a length of a space is changed , the phase of the transmitted light is unchanged ( for the default wavelength k = 0 ) , as shown the in the lower trace . 

so far we have considered the electromagnetic field to be monochromatic . this has allowed us to compute light - field amplitudes in a quasi - static optical setup . in this section ,
in fact , we consider a field consisting of a finite and discrete number of frequency components . we write this as 41\documentclass[12pt]{minimal } 
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				 \begin{document}$$e(t , z ) = \sum\limits_j { { a_j}\exp ( { \rm{i}}({\omega _ j}t - { k_j}z)),}$$\end{document } with complex amplitude factors aj , j as the angular frequency of the light field and kj = j / c . in many cases
the analysis compares different fields at one specific location only , in which case we can set z = 0 and write 42\documentclass[12pt]{minimal } 
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				 \begin{document}$$e(t ) = \sum\limits_j { { a_j}\exp ( { \rm{i}}{\omega _ j}t){.}}$$\end{document } in the following sections the concept of light modulation is introduced .
as this inherently involves light fields with multiple frequency components , it makes use of this type of field description .
again we start with the two - mirror cavity to illustrate how the concept of modulation can be used to model the effect of mirror motion .
laser interferometers typically use three different types of light fields : the laser with a frequency of , for example , f  2.8  10 hz , radio frequency ( rf ) sidebands used for interferometer control with frequencies ( offset to the laser frequency ) of f  1  10 to 150  10 hz , and the signal sidebands at frequencies of 1 to 10,000 hz3 . as these modulations
usually have as their origin a change in optical path length , they are often phase modulations of the laser frequency , the rf sidebands are utilised for optical readout purposes , while the signal sidebands carry the signal to be measured ( the gravitational - wave signal plus noise created in the interferometer ) .
figure 15 shows a time domain representation of an electromagnetic wave of frequency 0 , whose amplitude or phase is modulated at a frequency one can easily see some characteristics of these two types of modulation , for example , that amplitude modulation leaves the zero crossing of the wave unchanged whereas with phase modulation the maximum and minimum amplitude of the wave remains the same . in the frequency domain in which a modulated field is expanded into several unmodulated field components ,
the interpretation of modulation becomes even easier : any sinusoidal modulation of amplitude or phase generates new field components , which are shifted in frequency with respect to the initial field .
basically , light power is shifted from one frequency component , the carrier , to several others , the sidebands .
the relative amplitudes and phases of these sidebands differ for different types of modulation and different modulation strengths .
this section demonstrates how to compute the sideband components for amplitude , phase and frequency modulation . 

figure 15example traces for phase and amplitude modulation : the upper plot a ) shows a phase - modulated sine wave and the lower plot b ) depicts an amplitude - modulated sine wave .
phase modulation is characterised by the fact that it mostly affects the zero crossings of the sine wave .
the equations show the modulation terms in red with m the modulation index and  the modulation frequency .
example traces for phase and amplitude modulation : the upper plot a ) shows a phase - modulated sine wave and the lower plot b ) depicts an amplitude - modulated sine wave .
phase modulation is characterised by the fact that it mostly affects the zero crossings of the sine wave .
the equations show the modulation terms in red with m the modulation index and  the modulation frequency .
the number of sidebands with noticeable power depends on the modulation strength ( or depth ) given by the modulation index m. assuming an input field 43\documentclass[12pt]{minimal } 
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				 \begin{document}$${e_{{\rm{in } } } } = { e_0}\exp ( { \rm{i}}{\omega _ 0}t),$$\end{document } a sinusoidal phase modulation of the field can be described as 44\documentclass[12pt]{minimal } 
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				 \begin{document}$$e = { e_0}\exp \left({{\rm{i}}({\omega _ 0}t + m\cos ( \omega t ) ) } \right).$$\end{document } this equation can be expanded using the identity   45\documentclass[12pt]{minimal } 
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				 \begin{document}$$\exp ( { \rm{i}}z\cos \varphi ) = \sum\limits_{k = - \infty}^\infty { { { \rm{i}}^k}{j_k } } ( z)\exp ( { \rm{i}}\;k\varphi),$$\end{document } with bessel functions of the first kind jk(m )
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				 \begin{document}$$e = { e_0}\exp ( { \rm{i}}{\omega _ 0}t ) \sum\limits_{k = - \infty}^\infty { { { \rm{i}}^k}{j_k } } ( m)\exp ( { \rm{i}}\;k\omega t){.}$$\end{document } the field for k = 0 , oscillating with the frequency of the input field 0 , represents the carrier .
the sidebands can be divided into upper ( k > 0 ) and lower ( k < 0 ) sidebands .
these sidebands are light fields that have been shifted in frequency by k . the upper and lower sidebands with the same absolute value of k are called a pair of sidebands of order k. equation ( 46 ) shows that the carrier is surrounded by an infinite number of sidebands
. however , for small modulation indices ( m < 1 ) the bessel functions rapidly decrease with increasing k ( the lowest orders of the bessel functions are shown in figure 16 ) . for small modulation indices we can use the approximation   47\documentclass[12pt]{minimal } 
				
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				 \begin{document}$${j_k}(m ) = { \left({{m \over 2 } } \right)^k}\sum\limits_{n = 0}^\infty { { { { { \left({- { { { m^2 } } \over 4 } } \right)}^n } } \over { n!(k + n ) ! } } = { 1 \over { k ! } } } { \left({{m \over 2 } } \right)^k } + o({m^{k + 2}}).$$\end{document } in which case , only a few sidebands have to be taken into account . for m  1 we can write 48\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { e = { e_0}\exp ( { \rm{i}}{\omega _ 0}t)}\qquad \qquad \qquad \qquad
\quad\quad\quad\quad\quad\quad\quad \\ { \times \left({{j_0}(m ) - { \rm{i}}\;{j_{- 1}}(m)\;\exp ( - { \rm{i}}\omega t ) + { \rm{i}}\;{j_1}(m)\exp ( { \rm{i}}\omega t ) } \right),}\\ \end{array}$$\end{document } and with 49\documentclass[12pt]{minimal } 
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				 \begin{document}$${j_{- k}}(m ) = { ( - 1)^k}{j_k}(m),$$\end{document } we obtain 50\documentclass[12pt]{minimal } 
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				 \begin{document}$$e = { e_0}\exp ( { \rm{i}}{\omega _ 0}t)\left({1 + { \rm{i}}{m \over 2}\left({\exp ( - { \rm{i}}\omega t ) + \exp ( { \rm{i}}\omega t ) } \right ) } \right),$$\end{document } as the first - order approximation in m. in the above equation the carrier field remains unchanged by the modulation , therefore this approximation is not the most intuitive .
it is clearer if the approximation up to the second order in is given : 51\documentclass[12pt]{minimal } 
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				 \begin{document}$$e = { e_0}\exp ( { \rm{i}}{\omega _ 0}t)\left({1 - { { { m^2 } } \over 4 } + { \rm{i}}{m \over 2}\left({\exp ( - { \rm{i}}\omega t ) + \exp ( { \rm{i}}\omega t ) } \right ) } \right),$$\end{document } which shows that power is transferred from the carrier to the sideband fields
. 
 figure 16some of the lowest - order bessel functions jk(x ) of the first kind . for small
x the expansion shows a simple dependency and higher - order functions can often be neglected .
some of the lowest - order bessel functions jk(x ) of the first kind . for small
x the expansion shows a simple dependency and higher - order functions can often be neglected .
higher - order expansions in m can be performed simply by specifying the highest order of bessel function , which is to be used in the sum in equation ( 46 ) , i.e. , 52\documentclass[12pt]{minimal } 
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				 \begin{document}$$e = { e_0}\exp ( { \rm{i}}{\omega _ 0}t)\sum\limits_{k = - order}^{order } { { i^k}\;{j_k}(m)\exp ( { \rm{i}}\;k\omega t){.}}$$\end{document } for small modulation , indices , phase modulation and frequency modulation can be understood as different descriptions of the same effect .
following the same spirit as above we would assume a modulated frequency to be given by 53\documentclass[12pt]{minimal } 
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				 \begin{document}$$\omega = { \omega _ 0 } + { m^{\prime}}\cos ( \omega t),$$\end{document } and then we might be tempted to write 54\documentclass[12pt]{minimal } 
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				 \begin{document}$$e = { e_0}\;\exp \left({{\rm{i(}}{\omega _ 0 } + { m^{\prime}}\cos ( \omega t))t } \right),$$\end{document } which would be wrong . the frequency of a wave is actually defined as /(2 ) = f = d/dt .
thus , to obtain the frequency given in equation ( 53 ) , we need to have a phase of 55\documentclass[12pt]{minimal } 
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				 \begin{document}$${\omega _
0}t + { { m^{\prime } } \over \omega}\sin ( \omega t){.}$$\end{document } for consistency with the notation for phase modulation , we define the modulation index to be 56\documentclass[12pt]{minimal } 
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				 \begin{document}$$m = { { m^{\prime } } \over \omega } = { { \delta \omega } \over \omega},$$\end{document } with  as the frequency swing  how far the frequency is shifted by the modulation  and  the modulation frequency  how fast the frequency is shifted .
thus , a sinusoidal frequency modulation can be written as 57\documentclass[12pt]{minimal } 
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				 \begin{document}$$e = { e_0}\;\exp ( { \rm{i}}\varphi ) = { e_0}\;\exp \left({{\rm{i}}\left({{\omega _ 0}t + { { \delta \omega } \over \omega}\cos ( \omega t ) } \right ) } \right),$$\end{document } which is exactly the same expression as equation ( 44 ) for phase modulation .
the practical difference is the typical size of the modulation index , with phase modulation having a modulation index of m < 10 , while for frequency modulation , typical numbers might be m > 10 .
thus , in the case of frequency modulation , the approximations for small m are not valid .
the series expansion using bessel functions , as in equation ( 46 ) , can still be performed , however , very many terms of the resulting sum need to be taken into account . in contrast to phase modulation , ( sinusoidal ) amplitude modulation always generates exactly two sidebands . furthermore , a natural maximum modulation index exists : the modulation index is defined to be one ( m = 1 ) when the amplitude is modulated between zero and the amplitude of the unmodulated field .
if the amplitude modulation is performed by an active element , for example by modulating the current of a laser diode , the following equation can be used to describe the output field : 58\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { e = { e_0}\exp ( { \rm{i}}{\omega _ 0}t)\left({1 + m\cos ( \omega t ) } \right)}\qquad\qquad\qquad\qquad\\ { = { e_0}\exp ( { \rm{i}}{\omega _ 0}t)\left({1 + { m \over 2}\exp ( { \rm{i}}\omega t ) + { m \over 2}\exp ( - { \rm{i}}\omega t ) } \right)}\\ \end{array}.$$\end{document } however , passive amplitude modulators ( like acousto - optic modulators or electro - optic modulators with polarisers ) can only reduce the amplitude . in these cases ,
the following equation is more useful : 59\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { e = { e_0}\exp ( { \rm{i}}{\omega _ 0}t)\left({1 - { m \over 2}\left({1 - \cos ( \omega t ) } \right ) } \right)}\qquad\qquad\qquad\\ { \quad = { e_0}\exp ( { \rm{i}}{\omega _ 0}t)\left({1 - { m \over 2 } + { m \over 4}\exp ( { \rm{i}}\omega t ) + { m \over 4}\exp ( - { \rm{i}}\omega t ) } \right).}\\ \end{array}$$\end{document } a common method of visualising the behaviour of sideband fields in interferometers is to use phase diagrams in which each field amplitude is represented by an arrow in the complex plane .
we can think of the electric field amplitude e0 exp(i 0 t ) as a vector in the complex plane , rotating around the origin with angular velocity 0 . to illustrate or to help visualise the addition of several light fields it can be useful to look at this problem using a rotating reference frame , defined as follows .
a complex number shall be defined as z = x + iy so that the real part is plotted along the x - xis , while the y - axis is used for the imaginary part .
we want to construct a new coordinate system ( x , y ) in which the field vector is at a constant position .
this can be achieved by defining 60\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { x = x^{\prime}\;\cos { \omega _ 0}t - y^{\prime}\;\sin { \omega _ 0}t}\\ { y = x^{\prime}\;\sin { \omega _ 0}t + y^{\prime}\;\cos { \omega _ 0}t,}\\ \end{array}$$\end{document } or 61\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { x^{\prime } = x \cos ( - { \omega _ 0}t ) - y \sin ( - { \omega _ 0}t)}\\ { y^{\prime } = x \sin ( - { \omega _ 0}t ) + y \cos ( - { \omega _ 0}t){.}}\\ \end{array}$$\end{document } figure 17 illustrates how the transition into the rotating frame makes the field vector to appear stationary .
the angle of the field vector in a rotating frame depicts the phase offset of the field .
therefore these vectors are also called phasors and the illustrations using phasors are called phasor diagrams .
two more complex examples of how phasor diagrams can be employed is shown in figure 18 . 

figure 17electric field vector e0 exp(i0 t ) depicted in the complex plane and in a rotating frame ( x , y ) rotating at 0 so that the field vector appears stationary . 

the upper plots a ) illustrate how a phasor diagram can be used to describe phase modulation , while the lower plots b ) do the same for amplitude modulation . in both cases the left hand plot shows the carrier in blue and the modulation sidebands in green as snapshots at certain time intervals .
one can see clearly that the upper sideband ( 0 +  ) rotates faster than the carrier , while the lower sideband rotates slower .
the right plot in both cases shows how the total field vector at any given time can be constructed by adding the three field vectors of the carrier and sidebands .
[ drawing courtesy of simon chelkowski ] electric field vector e0 exp(i0 t ) depicted in the complex plane and in a rotating frame ( x , y ) rotating at 0 so that the field vector appears stationary . amplitude and phase modulation in the 
the upper plots a ) illustrate how a phasor diagram can be used to describe phase modulation , while the lower plots b ) do the same for amplitude modulation . in both cases the left hand plot shows the carrier in blue and the modulation sidebands in green as snapshots at certain time intervals .
one can see clearly that the upper sideband ( 0 +  ) rotates faster than the carrier , while the lower sideband rotates slower .
the right plot in both cases shows how the total field vector at any given time can be constructed by adding the three field vectors of the carrier and sidebands .
[ drawing courtesy of simon chelkowski ] phasor diagrams can be especially useful to see how frequency coupling of light field amplitudes can change the type of modulation , for example , to turn phase modulation into amplitude modulation .
mirror motion does not change the transmitted light ; however , the phase of the reflected light will be changed as shown in equation ( 11 ) .
the position modulation is given as xm = cos(st + s ) , and thus the reflected field at the mirror becomes ( assuming a4 = 0 ) 62\documentclass[12pt]{minimal } 
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				 \begin{document}$${a_3 } = r{a_1}\exp ( - { \rm{i\;2}}{\phi _ 0})\exp ( { \rm{i}}2k{x_{\rm{m } } } ) \approx r{a_1}\exp ( - { \rm{i\;2}}{\phi _ 0})\exp \left({- { \rm{i\;2}}{k_0}{a_{\rm{s}}}\cos ( { \omega _ { \rm{s}}}t + { \varphi _ { \rm{s } } } ) } \right),$$\end{document } setting m = 2k0as .
this can be expressed as 63\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { a_3 } = r{a_1}\exp ( - { \rm{i\;2}}{\phi _ 0})\left({1 + { \rm{i}}{m \over 2}\exp \left({- { \rm{i}}({\omega _ { \rm{s}}}t + { \varphi _ { \rm{s } } } ) } \right ) + { \rm{i}}{m \over 2}\exp \left({{\rm{i}}({\omega _ { \rm{s}}}t + { \varphi _ { \rm{s } } } ) } \right ) } \right)}\\ { = r{a_1}\exp ( - { \rm{i\;2}}{\phi _ 0})\left({1 + { m \over 2}\exp \left({- { \rm{i}}({\omega _ { \rm{s}}}t + { \varphi _ { \rm{s } } } - \pi/2 ) } \right ) } \right.}\qquad\qquad\qquad\\ { \left . { + { m \over 2}\exp \left({{\rm{i}}({\omega _ { \rm{s}}}t + { \varphi _ { \rm{s } } } + \pi/2 ) } \right ) } \right).}\qquad\qquad\qquad\qquad\qquad\qquad\\ \end{array}$$\end{document } 
 figure 19a sinusoidal signal with amplitude as frequency s and phase offset s is applied to a mirror position , or to be precise , to the mirror tuning .
the equation given for the tuning  assumes that s/0  1 , see section 2.5 .
a sinusoidal signal with amplitude as frequency s and phase offset s is applied to a mirror position , or to be precise , to the mirror tuning .
the equation given for the tuning  assumes that s/0  1 , see section 2.5 .
the coupling between electromagnetic fields at optical components introduced in section 2 referred only to the amplitude and phase of a simplified monochromatic field , ignoring all the other parameters of the electric field of the beam given in equation ( 1 ) .
however , this mathematical concept can be extended to include other parameters provided that we can find a way to describe the total electric field as a sum of components , each of which is characterised by a discrete value of the related parameters . in the case of the frequency of the light field , this means we have to describe the field as a sum of monochromatic components . in the previous sections we have shown how
this could be done in the special case of an initial monochromatic field that is subject to modulation : if the modulation index is small enough we can limit the amount of frequency components that we need to consider . in many cases it is actually sufficient to describe a modulation only by the interaction of the carrier at 0 ( the unmodulated field ) and two sidebands with a frequency offset of m to the carrier .
a beam given by the sum of three such components can be described by a complex vector : 64\documentclass[12pt]{minimal } 
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				 \begin{document}$$\vec a = \left({\begin{array}{*{20}c } { \quad a({\omega _ 0})}\\ { a({\omega _ 0 } - { \omega _ m})}\\ { a({\omega _ 0 } + { \omega _ m})}\\ \end{array } } \right ) = \left({\begin{array}{*{20}c } { { a_{\omega 0}}}\\ { { a_{\omega 1}}}\\ { { a_{\omega 2}}}\\ \end{array } } \right)$$\end{document } with 0 = 0 , 0  m = 1 and 0 + m = 2 . in the case of a phase modulator that applies a modulation of small modulation index m to an incoming light field \documentclass[12pt]{minimal } 
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				 \begin{document}${{\vec a}_1}$\end{document } , we can describe the coupling of the frequency component as follows : 65\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { a_{2,\omega 0 } } = { j_0}(m){a_{1,\omega 0 } } + { j_1}(m){a_{1,\omega 1 } } + { j_{- 1}}(m){a_{1,\omega 2}}}\;\;\\ { { a_{2,\omega 1 } } = { j_0}(m){a_{1,\omega 1 } } + { j_{- 1}}(m){a_{1,\omega 0}}}\qquad\qquad\qquad\\ { { a_{2,\omega 2 } } = { j_0}(m){a_{1,\omega 2 } } + { j_1}(m){a_{1,\omega 0}},}\qquad\qquad\qquad\\ \end{array}$$\end{document } which can be written in matrix form : 66\documentclass[12pt]{minimal } 
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				 \begin{document}$${\vec a_2 } = \left({\begin{array}{*{20}c } { { j_0}(m ) } & { { j_1}(m ) } & { { j_{- 1}}(m)}\\ { { j_{- 1}}(m ) } & { { j_0}(m ) } & 0\\ { { j_1}(m ) } & 0 & { { j_0}(m)}\\ \end{array } } \right){\vec a_1}.$$\end{document } and similarly , we can write the complete coupling matrix for the modulator component , for example , as 67\documentclass[12pt]{minimal } 
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				 \begin{document}$$\left({\begin{array}{*{20}c } { { a_{2,w0 } } } \\ { { a_{2,w1 } } } \\ { { a_{2,w2 } } } \\ { { a_{4,w0 } } } \\ { { a_{4,w1 } } } \\ { { a_{4,w2 } } } \\
\end{array } } \right)\left({\begin{array}{*{20}c } { { j_0}(m ) } & { { j_1}(m ) } & { { j_{- 1}}(m ) } & 0 & 0 & 0 \\ { { j_{- 1}}(m ) } & { { j_0}(m ) } & 0 & 0 & 0 & 0 \\ { { j_1}(m ) } & 0 & { { j_0}(m ) } & 0 & 0 & 0 \\ 0 & 0 & 0 & { { j_0}(m ) } & { { j_1}(m ) } & { { j_{- 1}}(m ) } \\ 0 & 0 & 0 & { { j_{- 1}}(m ) } & { { j_0}(m ) } & 0 \\ 0 & 0 & 0 & { { j_1}(m ) } & 0 & { { j_0}(m ) } \\ \end{array } } \right)\left({\begin{array}{*{20}c } { { a_{1,w0 } } } \\ { { a_{1,w1 } } } \\ { { a_{1,w2 } } } \\ { { a_{3,w0 } } } \\ { { a_{3,w1 } } } \\ { { a_{3,w2 } } }
phase modulation ( with up to five higher harmonics is applied to a laser beam and amplitude detectors are used to measure the field at the first three harmonics .
figure 20finesse example : modulation index.4 finesse example : modulation index.4 
 finesse input file for 
modulation index 

  finesse offers two different types of modulators : the  modulator component shown in the example above , and the  fsig command , which can be used to apply a signal modulation to existing optical components .
consequently finesse discards all nonlinear terms , which means that the sideband amplitude is proportional to the signal amplitude and harmonics are not created . 

in previous sections we have dealt with the amplitude of light fields directly and also used the amplitude detector in the finesse examples .
this is the advantage of a mathematical analysis versus experimental tests , in which only light intensity or light power can be measured directly .
the intensity of a field impinging on a photo detector is given as the magnitude of the poynting vector , with the poynting vector given as   68\documentclass[12pt]{minimal } 
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				 \begin{document}$$\vec s = \vec e \times \vec h = { 1 \over { { \mu _ 0}}}\vec e \times
\vec b.$$\end{document } inserting the electric and magnetic components of a plane wave , we obtain 69\documentclass[12pt]{minimal } 
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				 \begin{document}$$\vert \vec s\vert = { 1 \over { { \mu _ 0}c}}{e^2 } = c{\epsilon _ 0}e_0 ^ 2{\cos ^2}(\omega t ) = { { c{\epsilon _ 0 } } \over 2}e_0 ^ 2(1 + \cos ( 2\omega t)),$$\end{document } with 0 the electric permeability of vacuum and the speed of light . the response of a photo detector is given by the total flux of effective radiation4 during the response time of the detector .
for example , in a photo diode a photon will release a charge in the n - p junction .
the response time is given by the time it takes for the charge to travel through the detector ( and further time may be taken up in the electronic processing of the signal ) .
the size of the photodiode and the applied bias voltage determine the travel time of the charges with typical values of approximately 10 ns .
thus , frequency components faster than perhaps 100 mhz are not resolved by a standard photodiode .
for example , a laser beam with a wavelength of = 1064 nm has a frequency of f = c/  282 10 hz = 282 thz .
thus , the 2 component is much too fast for the photo detector ; instead , it returns the average power 70\documentclass[12pt]{minimal } 
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				 \begin{document}$$\vert \overline { \vec s } \vert = { { c{\epsilon _ 0 } } \over 2}e_0 ^ 2.$$\end{document } in complex notation we can write 71\documentclass[12pt]{minimal } 
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				 \begin{document}$$\vert \overline { \vec s } \vert = { { c{\epsilon _ 0 } } \over 2}ee^\ast.$$\end{document } however , for more intuitive results the light fields can be given in converted units , so that the light power can be computed as the square of the light field amplitudes . unless otherwise noted , throughout this work the unit of light field amplitudes \documentclass[12pt]{minimal } 
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				 \begin{document}$\sqrt { { \rm{watt}}}$\end{document}. thus , the notation used in this document to describe the computation of the light power of a laser beam is 72\documentclass[12pt]{minimal } 
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				 \begin{document}$$p = ee^\ast.$$\end{document } what is usually called an optical beat or simply a beat is the sinusoidal behaviour of the intensity of two overlapping and coherent fields .
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				 \begin{document}$$\begin{array}{*{20}c } { e = { e_0}\cos ( { \omega _ 1}t ) + { e_0}\cos ( { \omega _ 2}t)}\qquad\qquad\qquad\qquad\qquad\qquad\\ { p = { e^2 } = e_0 ^ 2\left({{{\cos}^2}({\omega _ 1}t ) + { { \cos}^2}({\omega _ 2}t ) + 2\cos ( { \omega _ 1}t)\cos ( { \omega _ 1}t ) } \right)}\;\;\;\;\\ { = e_0 ^ 2\left({{{\cos}^2}({\omega _
1}t ) + { { \cos}^2}({\omega _ 2}t ) + \cos ( { \omega _ + } t ) + \cos ( { \omega _ -}t ) } \right),}\\ \end{array}$$\end{document } with + = 1 + 2 and  = 1  2 . in this equation
the frequency  can be very small and can then be detected with the photodiode as illustrated in figure 22 .
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				 \begin{document}$${p_{{\rm{diode } } } } = e_0 ^ 2(1 + \cos ( { \omega _ -}t))$$\end{document } using the same example photodiode as before : in order to be able to detect an optical beat  would need to be smaller than 100 mhz . if we take two , sightly detuned nd : yag lasers with f = 282 thz , this means that the relative detuning of these lasers must be smaller than 10 . 

figure 22a beam with two frequency components hits the photo diode . shown in this plot
are the field amplitude , the corresponding intensity and the electrical output of the photodiode .
shown in this plot are the field amplitude , the corresponding intensity and the electrical output of the photodiode . in general , for a field with several frequency components , the photodiode signal can be written as 75\documentclass[12pt]{minimal } 
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				 \begin{document}$$\vert e{\vert ^2 } = e \cdot { e^{\ast } } = \sum\limits_{i = 0}^n { \sum\limits_{j = 0}^n { { a_i}a_j^{\ast}{e^{{\rm{i}}({\omega _ i}{\rm{-}}{\omega _ j})t}}.}}$$\end{document } for example , if the photodiode signal is filtered with a low - pass filter , such that only the dc part remains , we can compute the resulting signal by looking for all components without frequency dependence .
the frequency dependence vanishes when the frequency becomes zero , i.e. , in all parts of equation ( 75 ) with i = j .
the output is a real number , calculated like this : 76\documentclass[12pt]{minimal } 
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				 \begin{document}$$x = \sum\limits_i { \sum\limits_j { { a_i}a_j^{\ast}\quad { \rm{with}}\quad { \rm{\{}}i , j\vert i , j } \in \{0 , \ldots , n\ } \wedge { \omega _ i } = { \omega _ j}\ } { .}}$$\end{document } a typical application of light modulation , is its use in a modulation - demodulation scheme , which applies an electronic demodulation to a photodiode signal .
demodulation of a photodiode signal at a user - defined frequency x , performed by an electronic mixer and a low - pass filter , produces a signal , which is proportional to the amplitude of the photo current at dc and at the frequency interestingly , by using two mixers with different phase offsets one can also reconstruct the phase of the signal , or to be precise the phase difference of the light at 0  x with respect to the carrier light .
mathematically , the demodulation process can be described by a multiplication of the output with a cosine : cos(x+j ) ( is the demodulation phase ) , which is also called the  local oscillator. after the multiplication was performed only the dc part of the result is taken into account .
the signal is 77\documentclass[12pt]{minimal } 
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				 \begin{document}$${s_0 } = \vert e{\vert ^2 } = e \cdot e^\ast = \sum\limits_{i = 0}^n { \sum\limits_{j = 0}^n { { a_i}a_j^{\ast}{e^{{\rm{i}}({\omega _ i } - { \omega _ j})t}}.}}$$\end{document } multiplied with the local oscillator it becomes 78\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { s_1 } = { s_0 } \cdot \cos ( { \omega _ x}t + { \varphi _ x } ) = { s_0}{1 \over 2}({e^{{\rm{i}}({\omega _ x}t + { \varphi _ x } ) } } + { e^{{\rm{- i}}({\omega
_ x}t + { \varphi _ x } ) } } ) } \\ { = { 1 \over 2}\sum\limits_{i = 0}^n { \sum\limits_{j = 0}^n { { a_i}a_j^{\ast}{e^{{\rm{i}}({\omega _ i } - { \omega _ j})t } } \cdot ( { e^{{\rm{i}}({\omega _ x}t + { \varphi _ x } ) } } + { e^{{\rm{- i}}({\omega _ x}t + { \varphi _ x})}})}}. } \\
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				 \begin{document}${a_{ij } } = { a_i}a_j^{\ast}$\end{document } and \documentclass[12pt]{minimal } 
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				 \begin{document}${e^{{\rm{i}}{\omega _ { ij}}\,t } } = { e^{{\rm{i}}\,{{{\rm{(}}{\omega _ i}{\rm{-}}{\omega _ j})}^t}}}$\end{document } we can write 79\documentclass[12pt]{minimal } 
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				 \begin{document}$${s_1 } = { 1 \over 2}\left({\sum\limits_{i = 0}^n { { a_{ii } } + \sum\limits_{i = 0}^n { \sum\limits_{j = i + 1}^n { ( { a_{ij}}{e^{{\rm{i}}{\omega _ { ij}}t } } + a_{ij}^{\ast}{e^{{\rm{- i}}{\omega _ { ij}}t } } ) } } } } \right)\cdot({e^{{\rm{i}}({\omega
_ x}t + { \varphi _ x})}}){.}$$\end{document } when looking for the dc components of s1 we get the following : 80\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { s_{1,{\rm{dc } } } } = \sum\limits_{ij } { { 1 \over 2}({a_{ij}}\,{e^{- { \rm{i}}\,{\varphi _ x } } } + a_{ij}^{\ast } \,{e^{{\rm{i}}\,{\varphi _ x}}})\quad { \rm{with}}\quad \{i , j\vert i , j \in \{0 , \ldots , n\ } \wedge { \omega _ { ij } } = { \omega _ x}\ } } } \\ { = \sum\limits_{ij } { \re \{{a_{ij}}\,{e^{- { \rm{i}}\,{\varphi _ x}}}\ } { . } } \qquad \quad \qquad \qquad \qquad \qquad \qquad \qquad \qquad \,\ , } \\
\end{array}$$\end{document } this would be the output of a mixer and a subsequent low - pass filter .
the results for x = 0 and x = /2 are called in - phase and in - quadrature , respectively ( or also first and second quadrature ) . they are given by 81\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { s_{1,{\rm{dc , phase } } } } = \sum\limits_{ij } { \re \{{a_{ij}}\},}}\\ { { s_{1,{\rm{dc , quad } } } } = \sum\limits_{ij } { \im \{{a_{ij}}\}.}}\\\end{array}$$\end{document } if only one mixer is used , the output is always real and is determined by the demodulation phase .
however , with two mixers generating the in - phase and in - quadrature signals , it is possible to construct a complex number representing the signal amplitude and phase : 82\documentclass[12pt]{minimal } 
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				 \begin{document}$$z = \sum\limits_{ij } { { a_i}a_j^{\ast}\quad { \rm{with}}\quad \{i , j\vert i , j \in \{0 , \ldots , n\ } \wedge { \omega _ { ij } } = { \omega _ x}\}{.}}$$\end{document } often several sequential demodulations are applied in order to measure very specific phase information .
for example , a double demodulation can be described as two sequential multiplications of the signal with two local oscillators and taking the dc component of the result .
first looking at the whole signal , we can write : 83\documentclass[12pt]{minimal } 
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				 \begin{document}$${s_2 } = { s_0 } \cdot \cos ( { \omega _ x}t + { \varphi _ x})\cos ( { \omega _ y}t + { \varphi _ y}){.}$$\end{document } this can be written as 84\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { s_2 } = { s_0}{1 \over 2}(\cos ( { \omega
_ y}t + { \omega _ x}t + { \varphi _ y } + { \varphi _ x } ) + \cos ( { \omega _ y}t - { \omega _ x}t + { \varphi _ y } - { \varphi _ x}))}\\ { = { s_0}{1 \over 2}(\cos ( { \omega _ + } t + { \varphi _ + } ) + \cos ( { \omega
-})),}\qquad \qquad \qquad\quad \\ \end{array}$$\end{document } and thus reduced to two single demodulations . since we now only care for the dc component we can use the expression from above ( equation ( 82 ) ) .
these two demodulations give two complex numbers : 85\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { z1 = \sum\limits_{ij } { { a_{ij}}\quad { \rm{with}}\quad \{i , j\vert i , j \in \{0 , \ldots , n\ } \wedge { \omega _ i } - { \omega _ j } = { \omega _ + } \},}}\\ { z2 = \sum\limits_{ij } { { a_{kl}}\quad { \rm{with}}\quad \{k , l\vert k , l \in \{0 , \ldots , n\ } \wedge { \omega _ k } - { \omega _ l } = { \omega _ -}\}{.}}}\\\end{array}$$\end{document } the demodulation phases are applied as follows to get a real output ( two sequential mixers ) 86\documentclass[12pt]{minimal } 
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				 \begin{document}$$x = \re \{({z_1}\;{e^{- { \rm{i}}{\varphi _ x } } } + { z_2}\;{e^{{\rm{i}}{\varphi _ x}}}){e^{{\rm{- i}}{\varphi _ y}}}\}{.}$$\end{document } in a typical setup , a user - defined demodulation phase for the first frequency ( here is given .
if two mixers are used for the second demodulation , we can reconstruct the complex number 87\documentclass[12pt]{minimal } 
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				 \begin{document}$$z = { z_1}\;{e^{- { \rm{i}}{\varphi _ x } } } + { z_2}\;{e^{{\rm{i}}{\varphi _ x}}}.$$\end{document } more demodulations can also be reduced to single demodulations as above . in this example
the beams have a slightly different frequency : the second beam has a 10 khz offset with respect to the first ( and to the default laser frequency ) .
the plot illustrates the output of four different detectors in one of the beam splitter output ports , while the phase of the second beam is tuned from 0 to 180. the photodiode 
the amplitude detectors  ad1 and  ad10k detect the laser light at 0 hz ( default frequency ) and 10 khz respectively . both show a constant absolute \documentclass[12pt]{minimal } 
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				 \begin{document}$\sqrt { 1/2}$\end{document } and the detector 
ad10k tracks the tuning of the phase of the second laser beam . finally , the detector  pd10k resembles a photodiode with demodulation at 10 khz .
in fact , this represents a photodiode and two mixers used to reconstruct a complex number as shown in equation ( 82 ) .
one can see that the phase of the resulting electronic signal also directly follows the phase difference between the two laser beams . 

the large interferometric gravitational - wave detectors currently in operation are based on two fundamental interferometer topologies : the fabry - prot and the michelson interferometer .
the main instrument is very similar to the original interferometer concept used in the famous experiment by michelson and morley , published in 1887 .
the main difference is that modern instruments use laser light to illuminate the interferometer to achieve much higher accuracy .
already the first prototype by forward and weiss has thus achieved a sensitivity a million times better than michelson s original instrument .
in addition , in current gravitational - wave detectors , the michelson interferometer has been enhanced by resonant cavities , which in turn have been derived from the original idea for a spectroscopy standard published by fabry and prot in 1899 .
the following section will describe the fundamental properties of the fabry - prot interferometer and the michelson interferometer .
a thorough understanding of these basic instruments is essential for the study of the high - precision interferometers used for gravitational - wave detection .
we have computed the field amplitudes in a linear two - mirror cavity , also called fabry - prot interferometer , in section 2.2 . in order to understand the features of this optical instrument it is of interest to have a closer look at the power circulation in the cavity .
a typical optical layout is shown in figure 24 : two parallel mirrors form the fabry - prot cavity .
a laser beam is injected through the first mirror ( at normal incidence ) . 

figure 24typical optical layout of a two - mirror cavity , also called a fabry - prot interferometer .
two mirrors form the fabry - prot interferometer , a laser beam is injected through one of the mirrors and the reflected and transmitted light can be detected by photo detectors .
typical optical layout of a two - mirror cavity , also called a fabry - prot interferometer .
two mirrors form the fabry - prot interferometer , a laser beam is injected through one of the mirrors and the reflected and transmitted light can be detected by photo detectors .
the behaviour of the ( ideal ) cavity is determined by the length of the cavity l , the wavelength of the laser  and the reflectivity and transmittance of the mirrors .
assuming an input power of |a0| = 1 , we obtain 88\documentclass[12pt]{minimal } 
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				 \begin{document}$${p_1 } = \vert { a_1}{\vert ^2 } = { { { t_1 } } \over { 1 + { r_1}{r_2 } - 2{r_1}{r_2}\cos ( 2kl)}},$$\end{document } with k = 2/ , p , t = t and r = r , as defined in section 1.4 . similarly we could compute the transmission of the optical system as the input - output ratio of the field amplitudes .
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				 \begin{document}$${{{a_2 } } \over { { a_0 } } } = { { - { t_1}{t_2}\exp ( - { \rm{i}}\;kl ) } \over { 1 - { r_1}{r_2}\exp ( - { \rm{i}}\;2kl)}}$$\end{document } is the frequency - dependent transfer function of the cavity in transmission ( the frequency dependency is hidden inside the k = 2f / c ) . figure 25 shows a plot of the circulating light power i over the laser frequency .
the maximum power is reached when the cosine function in the denominator becomes equal to one , i.e. , at kl = n with n an integer .
the lowest power values are reached at anti - resonance when kl = ( n + 1/2). we can also rewrite 90\documentclass[12pt]{minimal } 
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				 \begin{document}$$2kl = \omega { { 2l } \over c } = 2\pi f{{2l } \over c } = { { 2\pi f } \over { { \rm{fsr}}}},$$\end{document } with fsr being the free - spectral range of the cavity as shown in figure 25 .
thus , it becomes clear that resonance is reached for laser frequencies 91\documentclass[12pt]{minimal } 
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				 \begin{document}$${f_r } = n \cdot { \rm{fsr}},$$\end{document } where n is an integer . 

figure 25power enhancement in a two - mirror cavity as a function of the laser - light frequency .
the peaks marks the resonances of the cavity , i.e. , modes of operation in which the injected light is resonantly enhanced .
the frequency distance between two peaks is called free - spectral range ( fsr ) .
power enhancement in a two - mirror cavity as a function of the laser - light frequency .
the peaks marks the resonances of the cavity , i.e. , modes of operation in which the injected light is resonantly enhanced .
the frequency distance between two peaks is called free - spectral range ( fsr ) .
another characteristic parameter of a cavity is its linewidth , usually given as full width at half maximum ( fwhm ) or its pole frequency , fp . in order to compute the linewidth we have to ask at which frequency
the circulating power becomes half the maximum : 92\documentclass[12pt]{minimal } 
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				 \begin{document}$$\vert { a_1}({f_p}){\vert ^2}\overset{!}{=}{1 \over 2}\vert { a_{1,\max}}\vert ^{2}.$$\end{document } this results in the following expression for the full linewidth : 93\documentclass[12pt]{minimal } 
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				 \begin{document}$${\rm{fwhm = 2}}{f_p } = { { 2{\rm{fsr } } } \over \pi}\arcsin \left({{{1 - { r_1}{r_2 } } \over { 2\sqrt { { r_1}{r_2 } } } } } \right).$$\end{document } the ratio of the linewidth and the free spectral range is called the finesse of a cavity : 94\documentclass[12pt]{minimal } 
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				 \begin{document}$$f = { { { \rm{fsr } } } \over { { \rm{fwhm}}}}={\pi \over { { \rm{2arcsin}}\left({{{1 - { r_1}{r_2 } } \over { 2\sqrt { { r_1}{r_2 } } } } } \right)}}.$$\end{document } in the case of high finesse , i.e. , r1 and r2 are close to 1 we can use the fact that the argument of the arcsin function is small and make the approximation 95\documentclass[12pt]{minimal } 
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				 \begin{document}$$f \approx { { \pi \sqrt { { r_1}{r_2 } } } \over { 1 - { r_1}{r_2 } } } \approx { \pi \over { 1 - { r_1}{r_2}}}.$$\end{document } the behaviour of a two mirror cavity depends on the length of the cavity ( with respect to the frequency of the laser ) and on the reflectivities of the mirrors . regarding the mirror parameters one distinguishes three cases5 : 
 when t1 < t2 the cavity is called undercoupledwhen t1 = t2 the cavity is called impedance matchedwhen t1 > t2 the cavity is called overcoupled when t1 < t2 the cavity is called undercoupled when t1 = t2 the cavity is called impedance matched when t1 > t2 the cavity is called overcoupled the differences between these three cases can seem subtle mathematically but have a strong impact on the application of cavities in laser systems . one of the main differences is the phase evolution of the light fields , which is shown in figure 26 .
the circulating power shows that the resonance effect is better used in over - coupled cavities ; this is illustrated in figure 27 , which shows the transmitted and circulating power for the three different cases . only in the impedance - matched case
given the same total transmission t1 + t2 , the overcoupled case allows for the largest circulating power and thus a stronger 
resonance effect of the cavity , for example , when the cavity is used as a mode filter .
figure 26this figure compares the fields reflected by , transmitted by and circulating in a fabry - prot cavity for the three different cases : over - coupled , under - coupled and impedance matched cavity ( in all cases t1 + t2 = 0.2 and the round - trip loss is 1% ) .
the traces show the phase and amplitude of the electric field as a function of laser frequency detuning . 

figure 27power transmitted and circulating in a two mirror cavity with input power 1 w. the mirror transmissions are set such that t1 + t2 = 0.8 and the reflectivities of both mirrors are set as r = 1  t. the cavity is undercoupled for t1 < 0.4 , impedance matched at t1 = t2 = 0.4 and overcoupled for t1 > 0.4 .
the transmission is maximised in the impedance - matched case and falls similarly for over or undercoupled settings .
however , the circulating power ( and any resonance performance of the cavity ) is much larger in the overcoupled case .
this figure compares the fields reflected by , transmitted by and circulating in a fabry - prot cavity for the three different cases : over - coupled , under - coupled and impedance matched cavity ( in all cases t1 + t2 = 0.2 and the round - trip loss is 1% ) .
the traces show the phase and amplitude of the electric field as a function of laser frequency detuning .
power transmitted and circulating in a two mirror cavity with input power 1 w. the mirror transmissions are set such that t1 + t2 = 0.8 and the reflectivities of both mirrors are set as r = 1  t. the cavity is undercoupled for t1 < 0.4 , impedance matched at t1 = t2 = 0.4 and overcoupled for t1 > 0.4 .
the transmission is maximised in the impedance - matched case and falls similarly for over or undercoupled settings .
however , the circulating power ( and any resonance performance of the cavity ) is much larger in the overcoupled case .
we came across the michelson interferometer in section 2.4 when we discussed the phase relation at a beam splitter .
the typical optical layout of the michelson interferometer is shown again in figure 28 : a laser beam is split by a beam splitter and send along two perpendicular interferometer arms .
the four directions seen from the beam splitter are called north , east , west and south .
the ends of these arms ( north and east ) are marked by highly reflective end mirrors , which reflect the beams back into themselves so that they can be recombined by the beam splitter .
generally , the michelson interferometer has two outputs , namely the so far unused beam splitter port ( south ) and the input port ( west ) .
both output ports can be used to obtain interferometer signals , however , most setups are designed such that the signals with high signal - to - noise ratios are detected in the south port . 

figure 28typical optical layout of a michelson interferometer : a laser beam is split into two and sent along two perpendicular interferometer arms .
we will label the directions in a michelson interferometer as north , east , west and south in the following .
the end mirrors reflect the beams such that they are recombined at the beam splitter .
the south and west ports of the beam splitter are possible output port , however , in many cases , only the south port is used .
typical optical layout of a michelson interferometer : a laser beam is split into two and sent along two perpendicular interferometer arms .
we will label the directions in a michelson interferometer as north , east , west and south in the following .
the end mirrors reflect the beams such that they are recombined at the beam splitter .
the south and west ports of the beam splitter are possible output port , however , in many cases , only the south port is used .
the michelson interferometer output is determined by the laser wavelength  , the reflectivity and transmittance of the beam splitter and the end mirrors , and the length of the interferometer arms . in many cases
the end mirrors are highly reflective and the beam splitter ideally a 50:50 beam splitter . in that case
, we can compute the output for a monochromatic field as shown in section 2.4 . using equation ( 20 )
we can write the field in the south port as 96\documentclass[12pt]{minimal } 
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				 \begin{document}$${e_s } = { e_0}{{\rm{i } } \over 2}({e^{{\rm{i}}2k{l_n } } } + { e^{{\rm{i}}2k{l_e}}}).$$\end{document } we define the common arm length and the arm - length difference as 97\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c}{\bar l = { { { l_n } + { l_e } } \over 2 } } \\
{ \delta l = { l_n } - { l_e},}\\\end{array}$$\end{document } which yield \documentclass[12pt]{minimal } 
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				 \begin{document}$2{l_n } = 2\bar l + \delta l$\end{document } and \documentclass[12pt]{minimal } 
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				 \begin{document}$2{l_e } = 2\bar l - \delta l$\end{document}. thus , we can further simplify to get 98\documentclass[12pt]{minimal } 
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				 \begin{document}$${e_s } = { e_0}{{\rm{i } } \over 2}{e^{{\rm{i}}2k\bar l}}({e^{{\rm{i}}\;k\delta l } } + { e^{{\rm{- i}}k\delta l } } ) = { e_0}\;{\rm{i}}{e^{{\rm{i}}2k\bar l}}\cos ( k\delta l){.}$$\end{document } the photo detector then produces a signal proportional to 99\documentclass[12pt]{minimal } 
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				 \begin{document}$$s = { e_s}e_s^{\ast } = { p_0}{\cos ^2}(k\delta l ) = { p_0}{\cos ^2}(2\pi \delta l/\lambda){.}$$\end{document } this signal is depicted in figure 29 ; it shows that the power in the south port changes between zero and the input power with a period of l/ = 0.5 .
the tuning at which the output power drops to zero is called the dark fringe .
current interferometric gravitational - wave detectors operate their michelson interferometer at or near the dark fringe . 

figure 29power in the south port of a symmetric michelson interferometer as a function of the arm length difference l .
power in the south port of a symmetric michelson interferometer as a function of the arm length difference l .
the above seems to indicate that the macroscopic arm - length difference plays no role in the michelson output signal .
however , this is only correct for a monochromatic laser beam with infinite coherence length . in real interferometers care
must be taken that the arm - length difference is well below the coherence length of the light source . in gravitational - wave detectors
the macroscopic arm - length difference is an important design feature ; it is kept very small in order to reduce coupling of laser noise into the output but needs to retain a finite size to allow the transfer of phase modulation sidebands from the input to the output port ; this is illustrated in the finesse example below and will be covered in detail in section 6.4 .
the power in the south port of a michelson detector varies as the cosine squared of the microscopic arm length difference .
the maximum output can be equal to the input power , but only if the michelson interferometer is symmetric and lossless .
the tuning for which the south port power is zero is referred to as the dark fringe . 

michelson power 

  this example demonstrates how a macroscopic arm length difference can cause different 
dark fringe tuning for injected fields with different frequencies . in this case , some of the 10 mhz modulation sidebands are transmitted when the interferometer is tuned to a dark fringe for the carrier light .
it is also the cause for transmission of laser noise ( especially frequency noise ) into the michelson output port when the interferometer is not perfectly symmetric . 

in this section we introduce interferometers as length sensing devices . in particular , we explain how the fabry - prot interferometer and the michelson interferometer can be used for high - precision measurements and that both require a careful control of the base length ( which is to be measured ) in order to yield their large sensitivity . in addition , we briefly introduce the general concepts of error signals and transfer functions , which are used to describe most essential features of length sensing and control . in general , we will call an error signal any measured signal suitable for stabilising a certain experimental parameter p with a servo loop .
the aim is to maintain the variable p at a user - defined value , the operating point , p0 .
therefore , the error signal must be a function of the parameter p. in most cases it is preferable to have a bipolar signal with a zero crossing at the operating point .
the slope of the error signal at the operating point is a measure of the  gain of the sensor ( which in the case of interferometers is a combination of optics and electronics ) .
transfer functions describe the propagation of a periodic signal through a plant and are usually given as plots of amplitude and phase over frequency . by definition
this means a transfer function is independent of the actual signal size . for small signals or small deviations ,
experimentally , network analysers are commonly used to measure a transfer function : one connects a periodic signal ( the source ) to an actuator of the plant ( which is to be analysed ) and to an input of the analyser .
a signal from a sensor that monitors a certain parameter of the plant is connected to the second analyser input . by mixing the source with the sensor signal
the analyser can determine the amplitude and phase of the input signal with respect to the source ( amplitude equals one and the phase equals zero when both signals are identical ) .
mathematically , transfer functions can be modeled similarly : applying a sinusoidal signal sin(st ) to the interferometer , e.g. , as a position modulation of a cavity mirror , will create phase modulation sidebands with a frequency offset of s to the carrier light .
if such light is detected in the right way by a photodiode , it will include a signal at the frequency component s , which can be extracted , for example , by means of demodulation ( see section 4.2 ) .
typically a transfer function of the error signal is required for the design of the respective electronic servo .
a  transfer function of the error signal usually refers to a very specific setup : the system is held at its operating point , such that , on average , \documentclass[12pt]{minimal } 
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				 \begin{document}$\bar p = { p_0}$\end{document}. a signal is applied to the system in the form of a very small sinusoidal disturbance of p. the transfer function is then constructed by computing for each signal frequency the ratio of the error signal and the injected signal .
the operating point shall be at 100\documentclass[12pt]{minimal } 
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				 \begin{document}$${x_{\rm{d } } } = 0\quad { \rm{and}}\quad { x_{{\rm{ep}}}}({x_{\rm{d } } } = 0 ) = 0$$\end{document } the optical transfer function \documentclass[12pt]{minimal } 
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				 \begin{document}${t_{{\rm{opt,}}{{\rm{x}}_{\rm{d}}}}}$\end{document } with respect to this error signal is defined by 101\documentclass[12pt]{minimal } 
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				 \begin{document}$${\tilde x_{{\rm{ep}}}}(f ) = { t_{{\rm{opt}},{{\rm{x}}_{\rm{d}}}}}{t_{\det}}{\tilde x_{\rm{d}}}(f),$$\end{document } with tdet as the transfer function of the sensor . in the following
, tdet is assumed to be unity . at the zero crossing the slope of the error signal represents the magnitude of the transfer function for low frequencies : 102\documentclass[12pt]{minimal } 
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				 \begin{document}$${\left\vert { { { d{x_{{\rm{ep } } } } } \over { d{x_{\rm{d } } } } } } \right\vert _ { \vert { x_{\rm{d } } } = 0 } } = \vert { t_{{\rm{opt}},{{\rm{x}}_{\rm{d}}}}}{\vert _ { \vert f \rightarrow 0}}$$\end{document } the quantity above will be called the error - signal slope in the following text . it is proportional to the optical gain |topt , xd| , which describes the amplification of the gravitational - wave signal by the optical instrument . 

figure 32example of an error signal : the top graph shows the electronic interferometer output signal as a function of mirror displacement .
the operating point is given as the zero crossing , and the error - signal slope is defined as the slope at the operating point .
the slope of the error signal ( left graph ) is equal to the low frequency limit of the transfer function magnitude ( see equation ( 102 ) ) .
example of an error signal : the top graph shows the electronic interferometer output signal as a function of mirror displacement .
the operating point is given as the zero crossing , and the error - signal slope is defined as the slope at the operating point . the right graph shows the magnitude of the transfer function mirror displacement error signal .
the slope of the error signal ( left graph ) is equal to the low frequency limit of the transfer function magnitude ( see equation ( 102 ) ) . in figure 25
we have plotted the circulating power in a fabry - prot cavity as a function of the laser frequency .
the steep features in this plot indicate that such a cavity can be used to measure changes in the laser frequency . from the equation for the circulating power ( see equation ( 88 ) ) , 103\documentclass[12pt]{minimal } 
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				 \begin{document}$${p_1}/{p_0 } = { { { t_1 } } \over { 1 + { r_1}{r_2 } - 2{r_1}{r_2}\cos ( 2kl ) } } = { { { t_1 } } \over d},$$\end{document } we can see that the actual frequency dependence is given by the cos(2kl ) term . writing this term as 104\documentclass[12pt]{minimal } 
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				 \begin{document}$$\cos ( 2kl ) = \cos \left({2\pi { { lf } \over c } } \right),$$\end{document } we can highlight the fact that the cavity is in fact a reference for the laser frequency in relation to the cavity length . if we know the cavity length very well , a cavity should be a good instrument to measure the frequency of a laser beam .
however , if we know the laser frequency very accurately , we can use an optical cavity to measure a length . in the following
we will detail the optical setup and behaviour of a cavity used for a length measurement .
if we make use of the resonant power enhancement of the cavity to measure the cavity length , we can derive the sensitivity of the cavity from the differentiation of equation ( 88 ) , which gives the slope of the trace shown in figure 25 , 105\documentclass[12pt]{minimal } 
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				 \begin{document}$${{d\;{p_1}/{p_0 } } \over { d\;l } } = { { - 4{t_1}{r_1}{r_2}k\sin ( 2kl ) } \over { { d^2}}},$$\end{document } with d as defined in equation ( 103 ) .
this is plotted in figure 33 together with the cavity power as a function of the cavity tuning . from figure 33
we can deduce a few key features of the cavity : 
 the cavity must be held as near as possible to the resonance for maximum sensitivity .
this is the reason that active servo control systems play an important role in modern laser interferometers.if we want to use the power directly as an error signal for the length , we can not use the cavity directly on resonance because there the optical gain is zero . a suitable error signal ( i.e. , a bipolar signal )
a control system utilising this method is often called dc - lock or offset - lock
. however , we show below that more elegant alternative methods for generating error signals exist.the differentiation of the cavity power looks like a perfect error signal for holding the cavity on resonance . a signal proportional to such differentiation
the cavity must be held as near as possible to the resonance for maximum sensitivity .
this is the reason that active servo control systems play an important role in modern laser interferometers .
if we want to use the power directly as an error signal for the length , we can not use the cavity directly on resonance because there the optical gain is zero .
a suitable error signal ( i.e. , a bipolar signal ) can be constructed by adding an offset to the light power signal .
a control system utilising this method is often called dc - lock or offset - lock
. however , we show below that more elegant alternative methods for generating error signals exist .
the differentiation of the cavity power looks like a perfect error signal for holding the cavity on resonance .
a signal proportional to such differentiation can be achieved with a modulation - demodulation technique .
a tuning of 360 refers to a change in the cavity length by one laser wavelength .
this illustrates that near resonance the cavity power changes very rapidly when the cavity length changes .
this scheme for stabilising the frequency of a light field to the length of a cavity , or vice versa , is based on much older techniques for performing very similar actions with microwaves and microwave resonators .
drever and hall have adapted such techniques for use in the optical regime   and today what is now called the pound - drever - hall technique can be found in a great number of different types of optical setups .
an example layout of this scheme is shown in figure 34 , in this case for generating a length ( or frequency ) signal of a two - mirror cavity .
the laser is passed through an electro - optical modulator , which applies a periodic phase modulation at a fixed frequency . in many cases
the modulation frequency is chosen such that it resides in the radio frequency band for which low - cost , low - noise electronic components are available .
however , from the frequency domain analysis introduced in section 5 , we know that in most cases not all the light can be injected into the cavity .
let s consider the example of an over - coupled cavity with the reflectivity of the end mirror r2 < 1 .
such a cavity would have a frequency response as shown in the top traces of figure 26 ( recall that the origin of the frequency axis refers to an arbitrarily chosen default frequency , which for this figure has been selected to be a resonance frequency of the cavity ) .
if the cavity is held on resonance for the unmodulated carrier field , this field enters the cavity , gets resonantly enhanced and a substantial fraction is transmitted .
if the frequency offset of the modulation sidebands is chosen such that it does not coincide with ( or is near to ) an integer multiple of the cavity s free spectral range , the modulation sidebands are mostly reflected by the cavity and will not be influenced as much by the resonance condition of the cavity as the carrier .
the photodiode measuring the reflected light will see the optical beat between the carrier field and the modulation sidebands .
this includes a component at the modulation frequency which is a measure of the phase difference between the carrier field and the sidebands ( given the setup as described above ) .
any slight change of the cavity length would introduce a proportional change in the phase of the carrier field and no change in the sideband fields .
thus the photodiode signal can be used to measure the length changes of the cavity .
one of the advantages of this method is the fact that the so - generated signal is bipolar with a zero crossing and steep slope exactly at the cavity s resonance , see figure 35 . 
 figure 34typical setup for using the pound - drever - hall scheme for length sensing and with a two - mirror cavity : the laser beam is phase modulated with an electro - optical modulator ( eom ) .
figure 35this figure shows an example of a pound - drever - hall ( pdh ) signal of a two - mirror cavity .
the plots refer to a setup in which the cavity mirrors are stationary and the frequency of the input laser is tuned linearly .
the three peaks correspond to the frequency tunings for which the carrier ( main central peak ) or the modulation sidebands ( smaller side peaks ) are resonant in the cavity .
coincident with the peaks in the upper trace are bipolar structures in the lower trace .
each of the bipolar structures would be suitable as a length - sensing signal . in most cases the central structure
is used , as experimentally it can be easily identified because its slope has a different sign compared to the sideband structures . typical setup for using the pound - drever - hall scheme for length
sensing and with a two - mirror cavity : the laser beam is phase modulated with an electro - optical modulator ( eom ) .
this figure shows an example of a pound - drever - hall ( pdh ) signal of a two - mirror cavity .
the plots refer to a setup in which the cavity mirrors are stationary and the frequency of the input laser is tuned linearly .
the three peaks correspond to the frequency tunings for which the carrier ( main central peak ) or the modulation sidebands ( smaller side peaks ) are resonant in the cavity .
coincident with the peaks in the upper trace are bipolar structures in the lower trace . each of the bipolar structures would be suitable as a length - sensing signal . in most cases the central structure
is used , as experimentally it can be easily identified because its slope has a different sign compared to the sideband structures . similarly to the two - mirror cavity , we can start to understand the length - sensing capabilities of the michelson interferometer by looking at the output light power as a function of a mirror movement , as shown in figure 29 .
the power changes as sine squared with the maximum slope at the point when the output power ( in what we call the south port ) is half the input power . the slope of the output power , which is the optical gain of the instrument for detecting a differential arm - length change l with a photo detector in the south port can be written as 106\documentclass[12pt]{minimal } 
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				 \begin{document}$${{ds } \over { d\delta l } } = { { 2\pi { p_0 } } \over \lambda}\sin \left({{{4\pi } \over \lambda}\delta l } \right)$$\end{document } and is shown in figure 36 .
the most notable difference of the optical gain of the michelson interferometer with respect to the fabry - prot interferometer ( see figure 33 ) is the wider , more smooth distribution of the gain .
this is due to the fact that the cavity example is based on a high - finesse cavity in which the optical resonance effect is dominant . in a basic michelson interferometer
the upper plot shows the output power of a michelson interferometer as detected in the south port ( as already shown in figure 29 ) .
the lower plot shows the optical gain of the instrument as given by the slope of the upper plot .
the upper plot shows the output power of a michelson interferometer as detected in the south port ( as already shown in figure 29 ) .
the lower plot shows the optical gain of the instrument as given by the slope of the upper plot .
however , the main difference is that the measurement is made differentially by comparing two lengths .
this allows one to separate a larger number of possible noise contributions , for example noise in the laser light source , such as amplitude or frequency noise .
this is why the main instrument for gravitational - wave measurements is a michelson interferometer .
however , the resonant enhancement of light power can be added to the michelson , for example , by using fabry - prot cavities within the michelson .
this construction of new topologies by combining michelson and fabry - prot interferometers will be described in detail in a future version of this review .
these can be represented by the positions ( along the optical axes ) of the end mirrors .
however , it is more efficient to use proper linear combinations of these and describe the michelson interferometer length or position information by the common and differential arm length , as introduced in equation ( 97 ) : \documentclass[12pt]{minimal } 
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\begin{document}$$\begin{array}{*{20}c}{\bar l = { { { l_n } + { l_e } } \over 2}}\quad\\{\delta l = { l_n } - { l_e}.}\\\end{array}$$\end{document } the michelson interferometer is intrinsically insensitive to the common arm length \documentclass[12pt]{minimal } 
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				 \begin{document}${\bar l}$\end{document}. similar to the fabry - prot cavity , the michelson interferometer is also often used to set an operating point where the optical gain of a direct light power detection is zero .
this operating point , given by l/ = ( 2n + 1 )  0.25 with n a non - negative integer , is called dark fringe .
this operating point has several advantages , the most important being the low ( ideally zero ) light power on the diode .
highly efficient and low - noise photodiodes usually use a small detector area and thus are typically not able to detect large power levels . by using the dark fringe operating point ,
the michelson interferometer can be used as a null instrument or null measurement , which generally is a good method to reduce systematic errors .
one approach to make use of the advantages of the dark fringe operating point is to use an operating point very close to the dark fringe at which the optical gain is not yet zero .
in such a scenario a careful trade - off calculation can be done by computing the signal - to - noise with noises that must be suppressed , such as the laser amplitude noise .
this type of operation is usually referred to as dc control or offset control and is very similar to the similarly - named mechanism used with fabry - prot cavities .
another option is to employ phase modulated light , similar to the pound - drever - hall scheme described in section 6.3 .
the optical layout of such a scheme is depicted in figure 37 : an electro - optical modulator is used to apply a phase modulation at a fixed ( usually rf type ) frequency to the ( monochromatic ) laser light before it enters the interferometer . the photodiode signal from the interferometer output
the method originally proposed by lise schnupp is also sometimes referred to as frontal modulation . 

figure 37this length sensing scheme is often referred to as frontal or schnupp modulation : an eom is used to phase modulate the laser beam before entering the michelson interferometer .
the signal of the photodiode in the south port is then demodulated at the same frequency used for the modulation .
this length sensing scheme is often referred to as frontal or schnupp modulation : an eom is used to phase modulate the laser beam before entering the michelson interferometer .
the signal of the photodiode in the south port is then demodulated at the same frequency used for the modulation .
the optical gain of a michelson interferometer with schnupp modulation is shown in figure 39 in section 6.6 . 

figure 33 shows a plot of the analytical functions describing the power inside a cavity and its differentiation by the cavity tuning .
finesse input file for  cavity power and slope 

  figure 39 shows the demodulated photodiode signal of a michelson interferometer with schnupp modulation , as well as its differentiation , the latter being the optical gain of the system . comparing this figure to figure 36
, it can be seen that with schnupp modulation , the optical gain at the dark fringe operating points is maximised and a suitable error signal for these points is obtained . 

in previous sections we have introduced a notation for describing the on - axis properties of electric fields .
specifically , we have described the electric fields along an optical axis as functions of frequency ( or time ) and the location z. models of optical systems may often use this approach for a basic analysis even though the respective experiments will always include fields with distinct off - axis beam shapes . a more detailed description of such optical systems needs to take the geometrical shape of the light field into account .
one method of treating the transverse beam geometry is to describe the spatial properties as a sum of  spatial components or  spatial modes so that the electric field can be written as a sum of the different frequency components and of the different spatial modes .
of course , the concept of modes is directly related to the use of a sort of oscillator , in this case the optical cavity .
most of the work presented here is based on the research on laser resonators reviewed originally by kogelnik and li .
siegman has written a very interesting historic review of the development of gaussian optics [ 52 , 51 ] and we use whenever possible the same notation as used in his textbook  lasers .
this section introduces the use of gaussian modes for describing the spatial properties along the transverse orthogonal x and y directions of an optical beam .
we can write 107\documentclass[12pt]{minimal } 
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				 \begin{document}$$e(t , x , y , z ) = \sum\limits_j { \sum\limits_{n , m } { { a_{jnm}}\;{u_{nm}}(x , y , z ) } \;\exp ( { \rm{i}}({\omega _ j}t - { k_j}z)),}$$\end{document } with unm as special functions describing the spatial properties of the beam and ajnm as complex amplitude factors ( j is again the angular frequency and kj = j / c ) . for simplicity
we restrict the following description to a single frequency component at one moment in time ( t = 0 ) , so 108\documentclass[12pt]{minimal } 
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				 \begin{document}$$e(x , y , z ) = \exp ( - { \rm{i}}\;kz)\;\sum\limits_{n , m } { { a_{nm } } { u_{nm}}(x , y , z){.}}$$\end{document } in general , different types of spatial modes unm can be used in this context .
of particular interest are the gaussian modes , which will be used throughout this document .
many lasers emit light that closely resembles a gaussian beam : the light mainly propagates along one axis , is well collimated around that axis and the cross section of the intensity perpendicular to the optical axis shows a gaussian distribution .
the following sections provide the basic mathematical framework for using gaussian modes for analysing optical systems .
mathematically , gaussian modes are solutions to the paraxial wave equation  a specific wave equation for electromagnetic fields .
all electromagnetic waves are solutions to the general wave equation , which in vacuum can be given as : 109\documentclass[12pt]{minimal } 
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				 \begin{document}$$\delta \vec e - { 1 \over { { c^2}}}\ddot \vec e = 0.$$\end{document } but laser light fields are special types of electromagnetic waves .
, a laser beam will have a characteristic length  describing the  width ( the dimension of the field transverse to the main propagation axis ) , and
a characteristic length l defining some local length along the propagation over which the beam characteristics do not vary much . by definition , for what we call a beam  is typically small and l large in comparison , so that /l can be considered small .
in fact , the paraxial wave equation ( and its solutions ) can be derived as the first - order terms of a series expansion of equation ( 109 ) into orders of /l . a simpler approach to
the paraxial - wave equation goes as follows : a particular beam shape shall be described by a function u(x , y , z ) so that we can write the electric field as 110\documentclass[12pt]{minimal } 
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				 \begin{document}$$e(x , y , z ) = u(x , y , z)\;\exp ( - { \rm{i}}\;kz){.}$$\end{document } substituting this into the standard wave equation yields a differential equation for u : 111\documentclass[12pt]{minimal } 
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				 \begin{document}$$(\partial _ x^2 + \partial
now we put the fact that u(x , y , z ) should be slowly varying with z in mathematical terms .
the variation of u(x , y , z ) with z should be small compared to its variation with x or y. also the second partial derivative in z should be small .
this can be expressed as 112\documentclass[12pt]{minimal } 
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				 \begin{document}$$\vert \partial _ z^2u(x , y , z)\vert \ll \vert 2k{\partial _ z}u(x , y , z)\vert , \vert \partial _ x^2u(x , y , z)\vert , \vert \partial _ y^2u(x , y , z)\vert.$$\end{document } with this approximation , equation ( 111 ) can be simplified to the paraxial wave equation , 113\documentclass[12pt]{minimal } 
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				 \begin{document}$$(\partial _ x^2 + \partial _
z}u(x , y , z ) = 0{.}$$\end{document } any field u that solves this equation represents a paraxial beam shape when used in the form given in equation ( 110 ) .
in general , any solution u(x , y , z ) of the paraxial wave equation , equation ( 113 ) , can be employed to represent the transverse properties of a scalar electric field representing a beam - like electromagnetic wave .
especially useful in this respect are special families or sets of functions that are solutions of the paraxial wave equation .
when such a set of functions is complete and countable , it s called a set of transverse electromagnetic modes ( tem ) .
for instance , the set of hermite - gauss modes are exact solutions of the paraxial wave equation .
the term complete means they can be understood as a base system of the function space defined by all solutions of the paraxial wave equation .
in other words , we can describe any solution of the paraxial wave equation u by a linear superposition of hermite - gauss modes : 114\documentclass[12pt]{minimal } 
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				 \begin{document}$$u^{\prime}(x , y , z ) = \sum\limits_{n , m } { { a_{jnm}}\;{u_{nm}}(x , y , z),}$$\end{document } which in turn allows us to describe any laser beam using a sum of these modes : 115\documentclass[12pt]{minimal } 
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				 \begin{document}$$e(t , x , y , z ) = \sum\limits_j { \sum\limits_{n , m } { { a_{jnm}}\;{u_{nm}}(x , y , z)\;\exp ( { \rm{i}}({\omega _ j}t - { k_j}z)).}}$$\end{document } the hermite - gauss modes as given in this document ( see section 7.5 ) are orthonormal so that 116\documentclass[12pt]{minimal } 
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				 \begin{document}$$\int \int dxdy\;{u_{nm}}u_{{n^\prime}{m^\prime}}^\ast = { \delta _ { n{n^\prime}}}{\delta _ { m{m^\prime } } } = \left\{{\begin{array}{*{20}c } { 1\quad { \rm{if}}\quad n = n^\prime\quad { \rm{and}}\quad m = m^\prime}\\ { 0\quad { \rm{otherwise}}\quad \quad \quad \quad \quad \quad}\\ \end{array } } \right\}.$$\end{document } this means that , in the function space defined by the paraxial wave equation , the hermite - gauss functions can be understood as a complete set of unit - length basis vectors .
, the power of a beam , as given by equation ( 108 ) , being detected on a single - element photodetector ( provided that the area of the detector is large with respect to the beam ) can be computed as 117\documentclass[12pt]{minimal } 
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				 \begin{document}$$ee^\ast = \sum\limits_{n , m } { { a_{nm}}a_{nm}^{\ast},}$$\end{document } or for a beam with several frequency components ( compare with equation ( 76 ) ) as 118\documentclass[12pt]{minimal } 
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				 \begin{document}$$ee^\ast = \sum\limits_{n , m } { \sum\limits_i { \sum\limits_j { { a_{inm}}a_{jnm\quad}^{\ast}{\rm{with}}\quad \{i , j\vert i , j \in \{0 , \ldots , n\ } } \wedge { \omega _ i } = { \omega _ j}\}}.}$$\end{document } the basic or  lowest - order hermite - gauss mode is equivalent to what is usually called a gaussian beam and is given by 119\documentclass[12pt]{minimal } 
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				 \begin{document}$$u(x , y , z ) = \sqrt { { 2 \over \pi } } { 1 \over { w(z)}}\exp ( { \rm{i}}\psi ( z))\;\exp \left({- { \rm{i}}\;k{{{x^2 } + { y^2 } } \over { 2{r_c}(z ) } } - { { { x^2 } + { y^2 } } \over { { w^2}(z ) } } } \right).$$\end{document } the parameters of this equation are explained in detail below .
the shape of a gaussian beam is quite simple : the beam has a circular cross section , and the radial intensity profile of a beam with total power p is given by 120\documentclass[12pt]{minimal } 
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				 \begin{document}$$i(r ) = { { 2p } \over { \pi { w^2}(z)}}\exp ( - 2{r^2}/{w^2}),$$\end{document } with  the spot size , defined as the radius at which the intensity is 1/e times the maximum intensity i(0 ) .
this is a gaussian distribution , see figure 40 , hence the name gaussian beam . 

the width of the beam is given by the radius  at which the intensity is 1/e of the maximum intensity . )
the width of the beam is given by the radius  at which the intensity is 1/e of the maximum intensity . )
figure 41 shows a different cross section through a gaussian beam : it plots the beam size as a function of the position on the optical axis . 

figure 41gaussian beam profile along z : this cross section along the x - z - plane illustrates how the beam size (z ) of the gaussian beam changes along the optical axis .
the position of minimum beam size 0 is called beam waist . see text for a description of the parameters  , zr and rc .
gaussian beam profile along z : this cross section along the x - z - plane illustrates how the beam size (z ) of the gaussian beam changes along the optical axis . the position of minimum beam size 0 is called beam waist .
see text for a description of the parameters  , zr and rc . such a beam profile ( for a beam with a given wavelength  ) can be completely determined by two parameters : the size of the minimum spot size 0 ( called beam waist ) and the position z0 of the beam waist along the z - axis . to characterise a gaussian beam ,
a gaussian beam can be divided into two different sections along the z - axis : a near field  a region around the beam waist , and a far field  far away from the waist .
the length of the near - field region is approximately given by the rayleigh range zr .
the rayleigh range and the spot size are related by 121\documentclass[12pt]{minimal } 
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				 \begin{document}$${z_{\rm{r } } } = { { \pi w_0 ^ 2 } \over \lambda}.$$\end{document } with the rayleigh range and the location of the beam waist , we can usefully write 122\documentclass[12pt]{minimal } 
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				 \begin{document}$$w(z ) = { w_0}\sqrt { 1 + { { \left({{{z - { z_0 } } \over { { z_{\rm{r } } } } } } \right)}^2}}.$$\end{document } this equation gives the size of the beam along the z - axis . in the far - field regime ( z  zr , z0 ) , it can be approximated by a linear equation , when 123\documentclass[12pt]{minimal } 
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				 \begin{document}$$w(z ) \approx { w_0}{z \over { { z_{\rm{r } } } } } = { { z\lambda } \over { \pi { w_0}}}.$$\end{document } the angle  between the z - axis and (z ) in the far field is called the diffraction angle6 and is defined by 124\documentclass[12pt]{minimal } 
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				 \begin{document}$$\theta = \arctan \left({{{{w_0 } } \over { { z_{\rm{r } } } } } } \right ) = \arctan \left({{\lambda \over { \pi { w_0 } } } } \right ) \approx { { { w_0 } } \over { { z_{\rm{r}}}}}.$$\end{document } another useful parameter is the radius of curvature of the wavefront at a given point z. the radius of curvature describes the curvature of the  phase front of the electromagnetic wave  a surface across the beam with equal phase  intersecting the optical axis at the position z. we obtain the radius of curvature as a function of z : 125\documentclass[12pt]{minimal } 
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				 \begin{document}$${r_c}(z ) = z - { z_0 } + { { z_{\rm{r}}^2 } \over { z - { z_0}}}.$$\end{document } we also find : 126\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { r_c } \approx \infty , \quad z - { z_0 } \ll { z_{\rm{r } } } } & { ( { \rm{beam}}\;{\rm{waist}})\quad \quad \quad \quad } \\ { { r_c } \approx z,\;\;\;\;\;\;\;z \gg { z_{\rm{r}}},{z_{\rm{0 } } } } & { ( { \rm{far}}\;{\rm{field}})\quad \quad \quad \quad \quad } \\ { { r_c } = 2{z_{\rm{r}}},\quad z - { z_0 } = { z_{\rm{r } } } } & { ( { \rm{maximum}}\;{\rm{curvature}})\;. } \\
\end{array}$$\end{document } if the interferometer is confined to a plane ( here the x  z plane ) , it is convenient to use projections of the three - dimensional description into two planes : the tangential plane , defined as the x  z plane and the sagittal plane as given by y and z. the beam parameters can then be split into two respective parameters : z0,s , 0,s for the sagittal plane and z0,t and 0,t for the tangential plane so that the hermite - gauss modes can be written as 127\documentclass[12pt]{minimal } 
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				 \begin{document}$${u_{nm}}(x , y ) = { u_n}(x,{z_{0,t}},{w_{0,t}})\;{u_m}(y,{z_{0,s}},{w_{0,s}}){.}$$\end{document } beams with different beam waist parameters for the sagittal and tangential plane are astigmatic .
this means one can use the separation into sagittal and tangential planes even if the actual optical system does not show this special type of symmetry .
, the term beam parameter generally refers to a simple case where 0,x = 0,y and z0,x = z0,y but all the results can also be applied directly to a pair of parameters .
the complete set of hermite - gauss modes is given by an infinite discrete set of modes unm(x , y , z ) with the indices n and m as mode numbers .
the term higher - order modes usually refers to modes with an order n + m > 0 .
the general expression for hermite - gauss modes can be given as   128\documentclass[12pt]{minimal } 
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				 \begin{document}$${u_{{\rm{nm}}}}(x , y , z ) = { u_{\rm{n}}}(x , z)\;{u_{\rm{m}}}(y , z),$$\end{document } with 129\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { u_{\rm{n}}}(x , z ) = { { \left({{2 \over \pi } } \right)}^{1/4}}{{\left({{{\exp ( { \rm{i}}(2n + 1)\psi ( z ) ) } \over { { 2^n}n!w(z ) } } } \right)}^{1/2 } } \times}\qquad\qquad\qquad\\ { { h_n}\left({{{\sqrt 2 x } \over { w(z ) } } } \right)\exp \left({- { \rm{i}}{{k{x^2 } } \over { 2{r_c}(z ) } } - { { { x^2 } } \over { { w^2}(z ) } } } \right),}\\ \end{array}$$\end{document } and hn(x ) the hermite polynomials of order n. the first hermite polynomials , without normalisation , can be written 130\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { h_0}(x ) = 1\quad \quad \;\;\;{h_1}(x ) = 2x}\qquad\;\\ { { h_2}(x ) = 4{x^2 } - 2\;{h_3}(x ) = 8{x^3 } - 12x.}\\ \end{array}$$\end{document } 
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				 \begin{document}$${h_{n + 1}}(x ) = 2x{h_n}(x ) - 2n{h_{n - 1}}(x).$$\end{document } further orders can be computed recursively since 132\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { u_{{\rm{nm}}}}(x , y , z ) = { { ( { 2^{n + m - 1}}n!m!\pi)}^{- 1/2}}{1 \over { w(z)}}\;\exp ( { \rm{i}}(n + m + 1)\psi ( z ) ) \times}\qquad \qquad \qquad \qquad\qquad \\ { { h_n}\left({{{\sqrt 2 x } \over { w(z ) } } } \right){h_m}\left({{{\sqrt 2 y } \over { w(z ) } } } \right)\exp \left({- { \rm{i}}{{k({x^2 } + { y^2 } ) } \over { 2{r_c}(z ) } } - { { { x^2 } + { y^2 } } \over { { w^2}(z ) } } } \right).}\\ \end{array}$$\end{document } for both transverse directions we can also rewrite the above to 133\documentclass[12pt]{minimal } 
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				 \begin{document}$${1 \over { q(z ) } } = { 1 \over { { r_c}(z ) } } - { \rm{i}}{\lambda \over { \pi { w^2}(z)}}.$$\end{document } the latter form has the advantage of clearly showing the extra phase shift along the z - axis of ( n + m + 1)(z ) called the gouy phase ; see section 7.8 . for a more compact description of the interaction of gaussian modes with optical components we will make use of the gaussian beam parameter q .
the beam parameter is a complex quantity defined as 134\documentclass[12pt]{minimal } 
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				 \begin{document}$$q(z ) = { \rm{i}}{z_{\rm{r } } } + z - { z_0 } = { q_0 } + z - { z_0}\quad { \rm{and}}\quad { q_0 } = { \rm{i}}{z_{\rm{r}}}.$$\end{document } it can also be written as 135\documentclass[12pt]{minimal } 
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				 \begin{document}$$u(x , y , z ) = { 1 \over { q(z)}}\exp \left({- { \rm{i}}\;k{{{x^2 } + { y^2 } } \over { 2q(z ) } } } \right).$$\end{document } using this parameter , equation ( 119 ) can be rewritten as 136\documentclass[12pt]{minimal } 
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				 \begin{document}$${w^2}(z ) = { \lambda \over \pi}{{\vert q{\vert ^2 } } \over { \im \{q\}}},$$\end{document } other parameters , like the beam size and radius of curvature , can also be written in terms of the beam parameter q : 137\documentclass[12pt]{minimal } 
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				 \begin{document}$$w_0 ^ 2 = { { \im \{q\ } \lambda } \over \pi},$$\end{document } 
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				 \begin{document}$${z_{\rm{r } } } = \im \{q\}$$\end{document } 
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				 \begin{document}$${r_c}(z ) = { { \vert q\vert ^{2 } } \over { \re \{q\}}}.$$\end{document } and 140\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { u_{{\rm{nm}}}}(x , y , z ) = { u_{\rm{n}}}(x , z){u_{\rm{m}}}(y , z)\quad { \rm{with}}}\qquad\qquad\qquad\qquad\qquad\qquad\qquad\qquad\quad \;\;\\ { u_{\rm n}(x , z ) = { { \left({{2 \over \pi } } \right)}^{1/4}}{{\left({{1 \over { { 2^n}n!{w_0 } } } } \right)}^{1/2}}{{\left({{{{q_0 } } \over { q(z ) } } } \right)}^{1/2}}{{\left({{{{q_0}\;q^\ast(z ) } \over { q_0^{\ast}\;q(z ) } } } \right)}^{n/2}}{h_n}\left({{{\sqrt 2 x } \over { w(z ) } } } \right)\exp \left({- { \rm{i}}{{k{x^2 } } \over { 2q(z ) } } } \right).}\\ \end{array}$$\end{document } the hermite - gauss modes can also be written using the gaussian beam parameter as7 

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				 \begin{document}$$\psi ( z ) = \arctan \left({{{z - { z_0 } } \over { { z_{\rm{r } } } } } } \right),$$\end{document } some of the properties of hermite - gauss modes can easily be described using cross sections of the field intensity or field amplitude .
figure 42 shows such cross sections , i.e. , the intensity in the x  y plane , for a number of higher - order modes .
this shows a x  y symmetry for mode indices n and m. we can also see how the size of the intensity distribution increases with the mode index , while the peak intensity decreases . 

one can see that the intensity distribution becomes wider for larger mode indices and the peak intensity decreases .
one can see that the intensity distribution becomes wider for larger mode indices and the peak intensity decreases .
similarly , figure 44 shows the amplitude and phase distribution of several higher - order hermite - gauss modes .
some further features of hermite - gauss modes : 
 the size of the intensity profile of any sum of hermite - gauss modes depends on z while its shape remains constant over propagation along the optical axis.the phase distribution of hermite - gauss modes shows the curvature ( or radius of curvature ) of the beam .
the size of the intensity profile of any sum of hermite - gauss modes depends on z while its shape remains constant over propagation along the optical axis .
the phase distribution of hermite - gauss modes shows the curvature ( or radius of curvature ) of the beam .
note that these are special features of gaussian beams and not generally true for arbitrary beam shapes .
figure 43 , for example , shows the amplitude and phase distribution of a triangular beam at the point where it is ( mathematically ) created and after a 10 m propagation .
figure 43these top plots show a triangular beam shape and phase distribution and the bottom plots the diffraction pattern of this beam after a propagation of z = 5 m. it can be seen that the shape of the triangular beam is not conserved and that the phase front is not spherical . 

figure 44these plots show the amplitude distribution and wave front ( phase distribution ) of hermite - gaussian modes unm ( labeled as hgnm in the plot ) .
all plots refer to a beam with  = 1 m ,  = 1 mm and distance to waist z = 1 m. the mode index ( in one direction ) defines the number of zero crossings ( along that axis ) in the amplitude distribution .
one can also see that the phase distribution is the same spherical distribution , regardless of the mode indices .
these top plots show a triangular beam shape and phase distribution and the bottom plots the diffraction pattern of this beam after a propagation of z = 5 m. it can be seen that the shape of the triangular beam is not conserved and that the phase front is not spherical .
these plots show the amplitude distribution and wave front ( phase distribution ) of hermite - gaussian modes unm ( labeled as hgnm in the plot ) .
all plots refer to a beam with  = 1 m ,  = 1 mm and distance to waist z = 1 m. the mode index ( in one direction ) defines the number of zero crossings ( along that axis ) in the amplitude distribution .
one can also see that the phase distribution is the same spherical distribution , regardless of the mode indices .
this gouy phase [ 8 , 26 , 25 ] describes the fact that , compared to a plane wave , the hermite - gauss modes have a slightly slower phase velocity , especially close to the waist .
the gouy phase can be written as 142\documentclass[12pt]{minimal } 
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				 \begin{document}$$\psi ( z ) = \arctan \left({{{\re \{q\ } } \over { \im \{q\ } } } } \right).$$\end{document } or , using the gaussian beam parameter , 143\documentclass[12pt]{minimal } 
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				 \begin{document}$$\varphi = ( n + m + 1)\psi ( z).$$\end{document } compared to a plane wave , the phase lag  of a hermite - gauss mode is 144\documentclass[12pt]{minimal } 
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				 \begin{document}$$\varphi = \left({n + { 1 \over 2 } } \right){\psi _ t}(z ) + \left({m + { 1 \over 2 } } \right){\psi _ s}(z),$$\end{document } with an astigmatic beam , i.e. , different beam parameters in the tangential and sagittal planes , this becomes 145\documentclass[12pt]{minimal } 
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				 \begin{document}$${\psi _ t}(z ) = \arctan \left({{{\re \{{q_t}\ } } \over { \im \{{q_t}\ } } } } \right),$$\end{document } with 146\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { u_{p , l}}(r , \phi , z ) = { 1 \over { w(z)}}\sqrt { { { 2p ! }
\over { \pi ( \vert l\vert + p ) ! } } } \exp ( { \rm{i}}(2p + \vert l\vert + 1)\psi ( z))}\qquad \qquad
\qquad\\ { \times \left({{{\sqrt 2 r } \over { w(z ) } } } \right)\;\;l_p^{(l)}\left({{{2{r^2 } } \over { w{{(z)}^2 } } } } \right)\exp \left({- { \rm{i}}\;k{{{r^2 } } \over { 2q(z ) } } + { \rm{i}}l\phi } \right),}\\ \end{array}$$\end{document } as the gouy phase in the tangential plane ( and is similarly defined in the sagittal plane ) .
laguerre - gauss modes are another complete set of functions , which solve the paraxial wave equation .
they are defined in cylindrical coordinates and can have advantages over hermite - gauss modes in the presence of cylindrical symmetry .
more recently , laguerre - gauss modes are being investigated in a different context : using a pure higher - order laguerre - gauss mode instead of the fundamental gaussian beam can significantly reduce the impact of mirror thermal noise on the sensitivity of gravitational wave detectors [ 54 , 12 ] .
laguerre - gauss modes are commonly given as   147\documentclass[12pt]{minimal } 
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				 \begin{document}$$l_p^{(l)}(x ) = { 1 \over { p!}}\sum\limits_{j = 0}^p { { { p ! } \over { j ! } } } \left({\begin{array}{*{20}c } { l + p } \\ { p - j } \\
\end{array } } \right){(- x)^j}.$$\end{document } with r ,  and z as the cylindrical coordinates around the optical axis .
the letter p is the radial mode index , l the azimuthal mode index8 and \documentclass[12pt]{minimal } 
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				 \begin{document}$l_p^{(l)}(x)$\end{document } are the associated laguerre polynomials : 148\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { u_{p , l}^{{\rm{alt}}}(r , \phi , z ) = { 2 \over { w(z)}}\sqrt { { { p ! } \over { ( 1 + { \delta _ { 0l}}\pi ( \vert l\vert + p ) ! } } } \exp ( { \rm{i}}(2p + \vert l\vert + 1)\psi ( z))}\qquad\quad\quad\\ { \times \left({{{\sqrt 2 r } \over { w(z ) } } } \right)\;\;l_p^{(l)}\left({{{2{r^2 } } \over { w{{(z)}^2 } } } } \right)\exp \left({- { \rm{i}}\;k{{{r^2 } } \over { 2q(z ) } } } \right)\cos ( l\phi){.}}\\\end{array}$$\end{document } all other parameters ( w(z ) , q(z ) ,  ) are defined as above for the hermite - gauss modes .
the dependence of the laguerre modes on  as given in equation ( 146 ) results in a spiraling phase front , while the intensity pattern will always show unbroken concentric rings ; see figure 45 .
these modes are also called helical laguerre - gauss modes because of the their special phase structure . 
 figure 45these plots show the amplitude distribution and wave front ( phase distribution ) of helical laguerre - gauss modes upl .
all plots refer to a beam with  = 1 pm ,  = 1 mm and distance to waist z = 1 m. these plots show the amplitude distribution and wave front ( phase distribution ) of helical laguerre - gauss modes upl .
all plots refer to a beam with  = 1 pm ,  = 1 mm and distance to waist z = 1 m. the reader might be more familiar with a slightly different type of laguerre modes ( compare figure 46 and figure 47 ) that features dark radial lines as well as dark concentric rings . mathematically , these can be described simply by replacing the phase factor exp(i l ) in equation ( 146 ) by a sine or cosine function .
for example , an alternative set of laguerre - gauss modes is given by   149\documentclass[12pt]{minimal } 
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				 \begin{document}$$u_{n , m}^{lg}(x , y , z ) = \sum\limits_{k = 0}^n { { { \rm{i}}^k}b(n , m , k)u_{n - k , k}^{hg}(x , y , z),}$$\end{document } this type of mode has a spherical phase front , just as the hermite - gauss modes .
we will refer to this set as sinusoidal laguerre - gauss modes throughout this document . 

the number of concentric dark rings is given by the radial mode index p. 
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				 \begin{document}$u_{pl}^{{\rm{alt}}}$\end{document}. the up0 modes are identical to the helical modes .
however , for azimuthal mode indices l > 0 the pattern shows l dark radial lines in addition to the p dark concentric rings .
the number of concentric dark rings is given by the radial mode index p. intensity profiles for sinusoidal laguerre - gauss modes \documentclass[12pt]{minimal } 
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				 \begin{document}$u_{pl}^{{\rm{alt}}}$\end{document}. the up0 modes are identical to the helical modes .
however , for azimuthal mode indices l > 0 the pattern shows l dark radial lines in addition to the p dark concentric rings . for the purposes of simulation it can be sometimes useful to decompose laguerre - gauss modes into hermite - gauss modes .
the mathematical conversion for helical modes is given as [ 7 , 1 ] 150\documentclass[12pt]{minimal } 
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				 \begin{document}$$b(n , m , k ) = \sqrt { { { ( n - k)!k ! } \over { { 2^n}n!m ! } } } { 1 \over { k!}}{({\partial _ t})^k}{[{(1 - t)^n}{(1 + t)^m}]_{t = 0}},$$\end{document } with real coefficients 151\documentclass[12pt]{minimal } 
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				 \begin{document}$$p_n^{\alpha , \beta}(x ) = { { { { ( - 1)}^n } } \over { { 2^n}n!}}{(1 - x)^{- \alpha}}{(1 + x)^{- \beta}}{({\partial _ x})^n}{(1 - x)^{\alpha + n}}{(1 + x)^{\beta + n}},$$\end{document } if n = n + m. this relates to the common definition of laguerre modes as upl as follows : p = min(n , m ) and l = n  m. the coefficients h(n , m , k ) can be computed numerically by using jacobi polynomials .
jacobi polynomials can be written in various forms : 152\documentclass[12pt]{minimal } 
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				 \begin{document}$$p_n^{\alpha , \beta}(x ) = { 1 \over { { 2^n}}}\sum\limits_{j = 0}^n { \left({\begin{array}{*{20}c } { n + \alpha}\\ { \;\;j}\\ \end{array } } \right)\left({\begin{array}{*{20}c } { n + \beta}\\ { n - j}\\ \end{array } } \right ) } { ( x - 1)^{n - j}}{(1 + x)^j},$$\end{document } or which leads to 153\documentclass[12pt]{minimal } 
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				 \begin{document}$$b(n , m , k ) = \sqrt { { { ( n - k)!k ! } \over { { 2^n}n!m ! } } } { ( - 2)^k}p_k^{n - k , m - k}(0).$$\end{document } whenever gauss modes are used to analyse an optical system , the gaussian beam parameters ( or equivalent waist sizes and locations ) must be defined for each location at which field amplitudes are to be computed ( or at which coupling equations are to be defined ) . in our experience
the quality of a computation or simulation and the correctness of the results depend critically on the choice of these beam parameters .
this is correct , but there is a practical limitation : the number of modes having non - negligible power might become very large if the beam parameters are not optimised , so that in practice a good set of beam parameters is usually required .
in general , the gaussian beam parameter of a mode is changed at every optical surface in a well - defined way ( see section 7.11 ) .
thus , a possible method of finding reasonable beam parameters for every location in the interferometer is to first set only some specific beam parameters at selected locations and then to derive the remaining beam parameters from these initial ones : usually it is sensible to assume that the beam at the laser source can be properly described by the ( hopefully known ) beam parameter of the laser s output mode .
in addition , in most stable cavities the light fields should be described by using the respective cavity eigenmodes .
then , the remaining beam parameters can be computed by tracing the beam through the optical system . 
trace in this context means that a beam starting at a location with an already - known beam parameter is propagated mathematically through the optical system . at every optical element along the path
the beam parameter is transformed according to the abcd matrix of the element ( see below ) .
the transformation of the beam parameter can be performed by the abcd matrix - formalism [ 34 , 50 ] .
when a beam passes an optical element or freely propagates though space , the initial beam parameter q1 is transformed into q2 .
this transformation can be described by four real coefficients as follows : 154\documentclass[12pt]{minimal } 
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				 \begin{document}$${{{q_2 } } \over { { n_2 } } } = { { a{{{q_1 } } \over { { n_1 } } } + b } \over { c{{{q_1 } } \over { { n_1 } } } + d}},$$\end{document } with the coefficient matrix 155\documentclass[12pt]{minimal } 
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				 \begin{document}$$m\left({\begin{array}{*{20}c } a & b\\ c & d\\ \end{array } } \right),$$\end{document } 
 n1 being the index of refraction at the beam segment defined by q1 , and n2 the index of refraction at the beam segment described by q2 .
abcd matrices for some common optical components are given below , for the sagittal and tangential plane .
a mirror in this context is a single , partly - reflecting surface with an angle of incidence of 90. the transmission is described by with rc being the radius of curvature of the spherical surface .
the sign of the radius is defined such that rc is negative if the centre of the sphere is located in the direction of propagation .
the curvature shown above ( in figure 48 ) , for example , is described by a positive radius .
figure 48 
 figure 49 the matrix for reflection is given by the reflection at the back surface can be described by the same type of matrix by setting c = 2n2/rc .
a beam splitter is understood as a single surface with an arbitrary angle of incidence 1 .
the matrices for transmission and reflection are different for the sagittal and tangential planes ( ms and mt ) : with 2 given by snell s law : 156\documentclass[12pt]{minimal } 
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				 \begin{document}$${n_1}\sin ( { \alpha _ 1 } ) = { n_2}\sin ( { \alpha _ 2}),$$\end{document } and n by 157\documentclass[12pt]{minimal } 
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				 \begin{document}$$\delta n = { { { n_2}\cos ( { \alpha _ 2 } ) - { n_1}\cos ( { \alpha _ 1 } ) } \over { \cos ( { \alpha _ 1})\cos ( { \alpha _ 2})}}.$$\end{document } if the direction of propagation is reversed , the matrix for the sagittal plane is identical and the matrix for the tangential plane can be obtained by changing the coefficients a and d as follows : 158\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { a \rightarrow 1/a , } \\ { d \rightarrow 1/d . } \\ \end{array}$$\end{document } 
 figure 50 the reflection at the front surface of a beam splitter is given by : to describe a reflection at the back surface the matrices have to be changed as follows : 159\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { r_c } \rightarrow - { r_c},}\\ { { n_1 } \rightarrow { n_2},}\quad\\ { { \alpha _ 1}\rightarrow - { \alpha _ 2}.}\\ \end{array}$$\end{document } 
 figure 51 a thin lens transforms the beam parameter as follows : where f is the focal length .
figure 52 as mentioned above , the beam in free space can be described by one base parameter q0 . in some cases
it is convenient to use a matrix similar to that used for the other components to describe the z - dependency of q(z ) = q0 + z. on propagation through a free space of the length l and index of refraction n , the beam parameter is transformed as follows .
in the plane - wave analysis section 1.4 , a laser beam is described in general by the sum of various frequency components of its electric field 160\documentclass[12pt]{minimal } 
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				 \begin{document}$$e(t , z ) = \sum\limits_j { { a_j}\;\exp \left({{\rm{i}}({\omega _ j}t - { k_j}z ) } \right).}$$\end{document } here we include the geometric shape of the beam by describing each frequency component as a sum of hermite - gauss modes : 161\documentclass[12pt]{minimal } 
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				 \begin{document}$$e(t , x , y , z ) = \sum\limits_j { \sum\limits_{n , m } { { a_{jnm}}\;{u_{nm}}(x , y)\;\exp ( { \rm{i}}({\omega _ j}t - { k_j}z))}.}$$\end{document } the shape of such a beam does not change along the z - axis ( in the paraxial approximation ) .
more precisely , the spot size and the position of the maximum intensity with respect to the z - axis may change , but the relative intensity distribution across the beam does not change its shape . each part of the sum may be treated as an independent field that can be described using the equation for plane - waves with only two exceptions : 
 the propagation through free space has to include the gouy phase shift , andupon reflection or transmission at a mirror or beam splitter the different hermite - gauss modes may be coupled ( see below ) . the propagation through free space has to include the gouy phase shift , and upon reflection or transmission at a mirror or beam splitter the different hermite - gauss modes may be coupled ( see below ) . the gouy phase shift can be included in the simulation in several ways .
for example , for reasons of flexibility the gouy phase has been included in finesse as a phase shift of the component space .
the first set is characterised by the beam parameter q1 and the second by the parameter q2 .
a beam with all power in the fundamental mode u00(q1 ) leaves the first cavity and is injected into the second . here , two 
misconfigurations are possible : 
 if the optical axes of the beam and the second cavity do not overlap perfectly , the setup is called misaligned , if the beam size or shape at the second cavity does not match the beam shape and size of the ( resonant ) fundamental eigenmode ( q1(zcav )  q2(zcav ) ) , the beam is then not mode - matched to the second cavity , i.e. , there is a mode mismatch .
if the optical axes of the beam and the second cavity do not overlap perfectly , the setup is called misaligned , if the beam size or shape at the second cavity does not match the beam shape and size of the ( resonant ) fundamental eigenmode ( q1(zcav )  q2(zcav ) ) , the beam is then not mode - matched to the second cavity , i.e. , there is a mode mismatch .
we consider the case in which the beam parameter for the input light is specified .
ideally , the abcd matrices then allow one to trace a beam through the optical system by computing the proper beam parameter for each beam segment . in this case
, the basis system of hermite - gauss modes is transformed in the same way as the beam , so that the modes are not coupled .
for example , an input beam described by the beam parameter q1 is passed through several optical components , and at each component the beam parameter is transformed according to the respective abcd matrix .
thus , the electric field in each beam segment is described by hermite - gauss modes based on different beam parameters , but the relative power between the hermite - gauss modes with different mode numbers remains constant , i.e. , a beam in a u00 mode is described as a pure u00 mode throughout the entire system . in practice , it is usually impossible to compute proper beam parameter for each beam segment as suggested above , especially when the beam passes a certain segment more than once . a simple case that illustrates this point is reflection at a spherical mirror .
we know that the proper beam parameter of the reflected beam is 162\documentclass[12pt]{minimal } 
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				 \begin{document}$${q_2 } = { { { q_1 } } \over { - 2{q_1}/{r_c } + 1}},$$\end{document } with rc being the radius of curvature of the mirror . in general , we get q1  q2 and thus two different  proper beam parameters for the same beam segment . only one special radius of curvature would result in matched beam parameters ( q1 = q2 ) .
hermite - gauss modes are coupled whenever a beam is not matched to a cavity or to a beam segment or if the beam and the segment are misaligned .
this coupling is sometimes referred to as  scattering into higher - order modes because in most cases the laser beam is a considered as a pure tem00 mode and any mode coupling would transfer power from the fundamental into higher - order modes .
however , in general , every mode with non - zero power will transfer energy into other modes whenever mismatch or misalignment occur , and this effect also includes the transfer from higher orders into a low order . to compute the amount of coupling the beam
must be projected into the base system of the cavity or beam segment it is being injected into .
this is always possible , provided that the paraxial approximation holds , because each set of hermite - gauss modes , defined by the beam parameter at a position z , forms a complete set .
such a change of the basis system results in a different distribution of light power in the new hermite - gauss modes and can be expressed by coupling coefficients that yield the change in the light amplitude and phase with respect to mode number .
let us assume that a beam described by the beam parameter q1 is injected into a segment described by the parameter q2 .
let the optical axis of the beam be misaligned : the coordinate system of the beam is given by ( x , y , z ) and the beam travels along the z - axis .
the beam segment is parallel to the z-axis and the coordinate system ( x , y , z ) is given by rotating the ( x , y , z ) system around the y - axis by the misalignment angle . the coupling coefficients are defined as 163\documentclass[12pt]{minimal } 
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				 \begin{document}$${u_{nm}}({q_1})\exp \left({{\rm{i}}(\omega t - kz ) } \right ) = \sum\limits_{n^{\prime},m^{\prime } } { { k_{n , m , n^{\prime},m^{\prime } } } } { u_{n^{\prime}m^{\prime}}}({q_2})\exp \left({{\rm{i}}(\omega t - kz^{\prime } ) } \right),$$\end{document } where unm(q1 ) are the hermite - gauss modes used to describe the injected beam and \documentclass[12pt]{minimal } 
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				 \begin{document}${u_{{n^{\prime}}\,{m^{\prime}}}}({q_2})$\end{document } are the 
note that including the plane wave phase propagation within the definition of coupling coefficients is very important because it results in coupling coefficients that are independent of the position on the optical axis for which the coupling coefficients are computed .
using the fact that the hermite - gauss modes unm are orthonormal , we can compute the coupling coefficients by the convolution   164\documentclass[12pt]{minimal } 
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				 \begin{document}$${k_{n , m , n^{\prime},m^{\prime } } } = \exp \left({{\rm{i}}2kz^{\prime}\sin^{2 } \left({{\gamma \over 2 } } \right ) } \right)\int { \int { dx^{\prime } } dy^{\prime } } \;{u_{n^{\prime}m^{\prime}}}\exp ( { \rm{i}}\;kx^{\prime}\sin \gamma)u_{nm}^{\ast}.$$\end{document } since the hermite - gauss modes can be separated with respect to x and y , the coupling coefficients can also be split into \documentclass[12pt]{minimal } 
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				 \begin{document}${k_{nm{n^{\prime}}{m^{\prime } } } } = { k_{n{n^{\prime}}}}{k_{m{m^{\prime}}}}$\end{document}. these equations are very useful in the paraxial approximation as the coupling coefficients decrease with large mode numbers . in order to be described as paraxial , the angle  must not be larger than the diffraction angle .
in addition , to obtain correct results with a finite number of modes the beam parameters q1 and q2 must not differ too much . the convolution given in equation ( 164 ) can be computed directly using numerical integration . however
another very good description of coupling coefficients and their derivation can be found in the work of vinet .
in   the above projection integral is partly solved and the coupling coefficients are given by simple sums as functions of  and the mode mismatch parameter k , which are defined by 165\documentclass[12pt]{minimal } 
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				 \begin{document}$$k = { 1 \over 2}({k_0 } + { \rm{i}}\;{k_2}),$$\end{document } where k0 = ( zr zr)/zr and k2 = ( ( z  z0 )  ( z  zr))/zrr .
this can also be written using q = i zr + z  z0 , as 166\documentclass[12pt]{minimal } 
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				 \begin{document}$$k = { { { \rm{i}}(q - q^{\prime})^\ast } \over { 2\im \{q^{\prime}\}}}.$$\end{document } the coupling coefficients for misalignment and mismatch ( but no lateral displacement ) can then be written as 167\documentclass[12pt]{minimal } 
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				 \begin{document}$${k_{nn^{\prime } } } = { ( - 1)^{n^{\prime}}}{e^{(x)}}{(n!n^{\prime}!)^{1/2}}{(1 + { k_0})^{n/2 + 1/4}}{(1 + k^\ast)^{- ( n + n^{\prime } + 1)/2}}\{{s_g } - { s_u}\},$$\end{document } where 168\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { { s_g } = \sum\limits_{\mu = 0}^{[n/2 ] } { \sum\limits_{\mu^{\prime } = 0}^{[n^{\prime}/2 ] } { { { { { ( - 1)}^\mu}{{\bar x}^{n - 2\mu}}{x^{n^{\prime}- 2\mu^{\prime } } } } \over { ( n - 2\mu)!(n^{\prime } - 2\mu^{\prime})!}}\sum\limits_{\sigma = 0}^{\min ( \mu , \mu^{\prime } ) } { { { { { ( - 1)}^\sigma}{{\bar f}^{\mu - \sigma}}{f^{\mu^{\prime } - \sigma } } } \over a}(2\sigma)!(\mu - \sigma)!(\mu^{\prime } - \sigma)!,}}}}\\ { { s_u } = \sum\limits_{\mu = 0}^{[(n - 1)/2 ] } { \sum\limits_{\mu^{\prime } = 0}^{[(n^{\prime } - 1)/2 ] } { { { { { ( - 1)}^\mu}{{\bar x}^{n - 2\mu - 1}}{x^{n^{\prime } - 2\mu^{\prime } - 1 } } } \over { ( n - 2\mu - 1)!(n^{\prime } - 2\mu^{\prime } - 1)!}}\sum\limits_{\sigma = 0}^{\min ( \mu , \mu^{\prime } ) } { { { { { ( - 1)}^\sigma}{{\bar f}^{\mu - \sigma}}{f^{\mu^{\prime } - \sigma } } } \over { ( 2\sigma + 1)!(\mu - \sigma)!(\mu^{\prime } - \sigma)!}}.}}}}\\ \end{array}$$\end{document } the corresponding formula for \documentclass[12pt]{minimal } 
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				 \begin{document}${k_{m{m^{\prime}}}}$\end{document } can be obtained by replacing the following parameters : n  m , n  m , x , \documentclass[12pt]{minimal } 
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				 \begin{document}$\bar x \rightarrow 0$\end{document } and e  1 ( see below ) .
the notation [ n/2 ] means 169\documentclass[12pt]{minimal } 
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				 \begin{document}$$\left [ { { m \over 2 } } \right ] = \left\{{\begin{array}{*{20}c } { m/2}\quad & { { \rm{if}}\;m\;{\rm{is}}\;{\rm{even } } , } \\ { ( m - 1)/2 } & { { \rm{if}}\;m\;{\rm{is}}\;{\rm{odd}}. } \\ \end{array } } \right.$$\end{document } the other abbreviations used in the above definition are 170\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { \bar x = ( { \rm{i}}z_{\rm{r}}^\prime - { z^\prime})\sin ( \gamma)/(\sqrt { 1 + k^\ast } { w_0}),}\\ { x = ( { \rm{i}}{z_{\rm{r } } } - { z^\prime})\sin ( \gamma)/(\sqrt { 1 + k^\ast } { w_0}),}\\ { f = k/(2(1 + { k_0}))},\\ { \bar f = k^\ast/2,}\\ { { e^{(x ) } } = \exp \left({- { { x\bar x } \over 2 } } \right).}\\ \end{array}$$\end{document } in general , the gaussian beam parameter might be different for the sagittal and tangential planes and a misalignment can be given for both possible axes ( around the y - axis and around the x - axis ) , in this case the coupling coefficients are given by 171\documentclass[12pt]{minimal } 
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				 \begin{document}$${k_{nmm^{\prime}n^{\prime } } } = { k_{nn^{\prime}}}{k_{mm^{\prime}}},$$\end{document } where \documentclass[12pt]{minimal } 
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				 \begin{document}${k_{n{n^{\prime}}}}$\end{document } is given above with 172\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { q \rightarrow { q_t}\quad \quad \quad}\\ { { \rm{and}}\quad \quad \quad \quad}\\ { { w_0 } \rightarrow { w_{t,0}},{\rm{etc}}.}\\ \end{array}$$\end{document } and   y is a rotation about the y - axis . the \documentclass[12pt]{minimal } 
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				 \begin{document}${k_{m{m^{\prime}}}}$\end{document } can be obtained with the same formula , with the following substitutions : 173\documentclass[12pt]{minimal } 
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				 \begin{document}$$\begin{array}{*{20}c } { n \rightarrow m,\quad \quad \quad}\\ { { n^\prime } \rightarrow { m^\prime},\quad \quad \quad}\\ { q \rightarrow { q_s},\quad \quad \quad}\\ { { \rm{thus}}\quad \quad \quad \quad \;}\\ { { w_0 } \rightarrow { w_{s,0}},{\rm{etc}}.}\\\end{array}$$\end{document } and   x is a rotation about the x - axis . at each component
a matrix of coupling coefficients has to be computed for transmission and reflection ; see figure 54 . 
 figure 54coupling coefficients for hermite - gauss modes : for each optical element and each direction of propagation complex coefficients k for transmission and reflection have to be computed . in this figure k1 ,
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				 \begin{document}${k_{nm{n^{\prime}}{m^{\prime}}}}$\end{document } describing the coupling of un , m into \documentclass[12pt]{minimal } 
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				 \begin{document}${u_{{n^{\prime}},{m^{\prime}}}}$\end{document}. coupling coefficients for hermite - gauss modes : for each optical element and each direction of propagation complex coefficients k for transmission and reflection have to be computed . in this figure k1 , k2 , k3 ,
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				 \begin{document}${k_{nm{n^{\prime}}{m^{\prime}}}}$\end{document } describing the coupling of un , m into \documentclass[12pt]{minimal } 
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				 \begin{document}${u_{{n^{\prime}},{m^{\prime}}}}$\end{document}. this example illustrates a possible use of the beam parameter detector 
bp : the beam radius of the laser beam is plotted as a function of distance to the laser . for this simulation
figure 55finesse example : beam parameter finesse example : beam parameter 
 finesse input file for  beam parameter 

  this example uses the 
tem command to create a laser beam which is a sum of equal parts in u00 and u10 modes .
this beam is passed through a triangular cavity , which acts as a mode cleaner .
being resonant for the u00 , the cavity transmits this mode and reflects the u10 mode as can be seen in the resulting plots . 

finesse input file for  mode cleaner 


 figure 56finesse example : mode cleaner 
  finesse example : mode cleaner finesse uses the hermite - gauss modes as a base system for describing the spatial properties of laser beams .
however , laguerre - gauss modes can be created using the coefficients given in equation ( 149 ) .
the ring structure in the phase plot is due to phase jumps , which could be removed by applying a phase 
the ring structure in the phase plot is due to phase jumps , which could be removed by applying a phase 
 several km - scale gravitational - wave detectors have been constructed world wide . 
 these instruments combine a number of advanced technologies to push the limits of precision length measurement . 
 the core devices are laser interferometers of a new kind ; developed from the classical michelson topology these interferometers integrate additional optical elements , which significantly change the properties of the optical system . 
 much of the design and analysis of these laser interferometers can be performed using well - known classical optical techniques , however , the complex optical layouts provide a new challenge . in this review 
 we give a textbook - style introduction to the optical science required for the understanding of modern gravitational wave detectors , as well as other high - precision laser interferometers . 
 in addition , we provide a number of examples for a freely available interferometer simulation software and encourage the reader to use these examples to gain hands - on experience with the discussed optical methods . 
Introduction
Optical Components: Coupling of Field Amplitudes
Light with Multiple Frequency Components
Optical Readout
Basic Interferometers
Interferometric Length Sensing and Control
Beam Shapes: Beyond the Plane Wave Approximation
Interferometer Matrix with Hermite-Gauss Modes