Power-supply circuits, such as for example AC-to-DC or DC-to-DC switching power supplies, are well known in the art.
FIG. 1 shows an architecture of a power-supply circuit that supplies at output a supply signal for a load LD.
In the example considered, the power-supply circuit comprises an input stage 10, a switching stage 20, an output stage 30, and a control circuit 40.
For instance, the input stage 10 may comprise a rectifier, such as for example a diode bridge, and/or one or more input filters. For instance, frequently the input stage 10 is configured to receive an input AC or DC voltage, for example via the electrical line M, and supplying at output a DC voltage Vin. In general, in particular when the input voltage M is already a DC voltage, the above filters may also be superfluous, and consequently the input stage 10 is purely optional.
The switching stage 20 consists of an electronic converter comprising at least one electronic switch. There exist many types of electronic converters that are divided mainly into insulated converters and non-insulated converters. For instance, non-insulated electronic converters are converters of the “buck”, “boost”, “buck-boost”, “Cuk”, “SEPIC” and “ZETA” type. Instead, insulated converters are, for example, converters of the “flyback”, “forward”, “half-bridge”, and “full-bridge” type. These types of converters are well known to the person skilled in the art.
Finally, the output stage 30 may comprise filters that stabilize the signal Vout at output from the switching stage 20. In general, these filters may also be included already in the stage 20, and consequently the output stage 30 is purely optional.
In the above architecture, switching of the switch or switches of the switching stage 20 is usually controlled via a control circuit 40, which opens and closes via at least one driving signal DRV for driving the switch or switches of the switching stage 20 as a function of at least one control signal. In general, there may be used:
a) an open-loop control (or forward, or predictive, or feed-forward control) via a control signal FF picked up, for example, on the input of block 10 or block 20; and/or
b) a closed-loop control (or feedback, or backward, control) via a control signal FB picked up, for example, on the output of block 20 or block 30.
For instance, illustrated in FIG. 1 is a feedback of the supply signal at output from block 20, such as for example the output voltage or current. Consequently, in this case, the control circuit 40 can drive the switch or switches of the switching stage 20 in such a way as to reach a desired output voltage or current.
For instance, FIG. 2 illustrates the circuit diagram of a flyback converter that can be used in the stage 20.
As is well known, a flyback converter comprises a transformer T with a primary winding T1 and a secondary winding T2, an electronic switch 204, such as for example an n-channel MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor) or a bipolar transistor or an IGBT (Insulated-Gate Bipolar Transistor), an output diode Dout, and an output capacitor Cout.
In particular, the transformer T can be modelled as an inductor Lm connected in parallel with the primary winding T1, which represents the magnetization inductance of the transformer T, an inductor Lr connected in series with the secondary winding T2, which represents the dispersion inductance of the transformer T, and an ideal transformer with a given turns ratio 1:n.
In the example considered, the converter 20 receives at input, through two input terminals 202 and GND1, a voltage Vin and supplies at output, through two output terminals 206 and GND2, a voltage Vout and a current iout.
As mentioned previously, the voltage Vin can be obtained also from an alternating current at input, for example via the input stage 10, which comprises a rectifier, such as for example a diode or a diode bridge and possibly one or more filters, such as for example capacitors.
In particular, the first input terminal 202 is connected to the first terminal of the primary winding T1 of the transformer T, and the second input terminal GND1 represents a first ground. Instead, the second terminal of the primary winding T1 of the transformer T is connected through the switch 204 to ground GND1. Consequently, the switch 204 can be used for selectively activating the flow of current through the primary winding T1 of the transformer T.
Instead, the first terminal of the secondary winding T2 of the transformer T is connected through a diode Dout to the first output terminal 208, and the second terminal of the secondary winding T2 of the transformer T is directly connected to a second output terminal GND2 that represents a second ground, which, on account of the insulating effect of the transformer T, is preferably different from the ground GND1 and is consequently represented by a different ground symbol. In general, it is sufficient for the secondary winding T2 and the diode Dout to be connected in series between the terminal 206 and the ground GND2.
Finally, an output capacitor Cout is connected in parallel with the output, i.e., between the terminals 206 and GND2.
Consequently, when the switch 204 is closed, the primary winding T1 of the transformer T is directly connected to the input voltage Vin. This results in an increase in the magnetic flux in the transformer T. Consequently, the voltage across the secondary winding T2 is negative, and the diode Dout is reverse biased. In this condition, the output capacitor Cout supplies the energy required by the load.
Instead, when the switch 204 is open, the energy stored in the transformer T is transferred as flyback current to the load.
As mentioned previously, the control may be in current or in voltage. For this purpose, a control unit 40 is typically used, which drives the switch 204 in such a way that the output voltage Vout or the output current iout is regulated on a desired value. For this purpose, a sensor configured for detecting the current iout or the voltage Vout may be used in a way in itself known.
Typically, the control unit 40 drives the switch 204 with a modulation of a PWM (Pulse-Width Modulation) type, in which the switch 204 is closed during a first operating interval, and the switch 204 is opened during a second operating interval. The person skilled in the art will appreciate that this PWM driving and control of the duration of the operating intervals are well known and may be obtained, for example, via a feedback of the voltage or of the current at output through an error amplifier. For instance, in the case of a current control, the duration of the first interval is increased until the (mean) current at output reaches a predetermined threshold.
With a PWM driving of this sort, there typically exist three operating modes. In particular, if the current in the magnetization inductance Lm never reaches zero during a switching cycle, the converter is said to be operating in CCM (Continuous-Current Mode). Instead, when the current in the magnetization inductance Lm reaches zero during the period, the converter is said to be operating in DCM (Discontinuous-Current Mode). Typically, the converter operates in discontinuous-current mode when the load absorbs a low current, and in continuous-current mode at higher levels of current absorption. The limit between CCM and DCM is reached when the current reaches zero exactly at the end of the switching cycle. This limit case is referred to as “TM” (Transition Mode). Furthermore, there exists the possibility of driving the switch also with a variable switching frequency, such as for example a resonant or quasi-resonant driving, where the switch 204 is switched when the voltage across the electronic switch 204 is zero or reaches a local minimum. Typically, the switching frequency, i.e., the sum of the durations of the operating periods, is fixed for CCM or DCM driving and variable for quasi-resonant driving.
A problem of these switching power-supply circuits is linked to the electronic consumption of the various components.
For instance, typically the control circuit 40 must always remain turned on for detecting the control signals FF and/or FB and for driving the switching stage 20.
However, at low loads, for example in the absence of loads connected to the converter, the feedback signal or signals FB may change even slowly. For this reason, the energy consumption of the control circuit 40 (and of the entire converter) can be reduced by activating and deactivating the control circuit 40 for certain periods. For instance, the control circuit 40 can be set in an energy-saving mode, the so-called “stand-by mode”, and the control circuit 40 can be reactivated periodically and/or as a function of a control signal. Consequently, in this operating mode, the switching stage 20 is not always driven, but switching of the switch or switches of the switching stage 20 is intermittent, and consequently this mode is typically referred to as “burst mode”.
For instance, in the sector of switching power supplies with galvanic insulation between the output voltage and the input voltage, the control feedback is usually obtained by means of an optocoupler device, which, in addition to closing the control loop, enables precisely galvanic insulation to be obtained. The advantage of this solution lies in the fact that the frequency of activation of the control circuit 40 and of the stage 20 depends upon the load of the system. However, frequently this solution is inefficient from the standpoint of consumption in stand-by conditions since the consumption of the feedback network of the optocoupler cannot be eliminated.
Other techniques enable execution of the feedback of the output voltage directly from the primary winding, without the aid of an optocoupler. In these systems, in conditions of zero load, the minimum frequency of the burst mode is typically fixed by the device and is a fixed frequency. In these systems, the switch or switches of the stage 20 must be turned back on periodically in order to transfer to the primary winding the information regarding the value of output voltage.
In particular, when the system is turned back on, it supplies at output a fixed energy that has to be dissipated in order to prevent the system from going out of regulation in the case of very low or zero loads. In order to overcome this problem, frequently a dummy load is inserted at output. The energy to be dissipated mainly depends upon the turning-on frequency, which should not be chosen low at will since between one reactivation of switching and the next the system is “blind”; i.e., there is no information on the state of the output. Once switching has taken place, the system can recognize a variation of the load and respond by supplying the necessary energy. In the worst case, i.e., variation of load from zero to the maximum value, the current absorbed by the load is sustained by the output capacitance, and the voltage drop Vout depends upon the value of this capacitance (the higher it is, the lower the voltage drop), upon the turning-on frequency (low frequencies result in high voltage drops), and upon the maximum current that can be applied at output.
For this reason, it is necessary to establish, in the design stage, a trade-off between consumption in stand-by conditions and the value of the output capacitance. For instance, to achieve dissipation of powers lower 5 mW there is usually required a turning-on period longer than 4 ms, which results in the use of output capacitances of the order of microfarads.
An example of implementation of this control technique from the primary is described, for instance, in the U.S. Pat. No. 6,590,789 (incorporated by reference).
Like classic insulated switching converters with feedback control obtained by means of an optocoupler, also those with feedback obtained by means of a primary winding consequently present considerable limits as regards obtaining high performance in terms of consumption levels in stand-by or zero-load conditions.
A way to overcome the above problem is to provide a system on the secondary, which, in the burst phase, monitors the output voltage Vout and, when this drops below a certain threshold, “wakes up”, by means of an appropriate communication mechanism and wake-up signal, the primary device. In this way, it is possible to obtain low dissipation without the use of high output capacitances.
Since the controller on the secondary is supplied by the output voltage of the converter, it cannot be fully enabled if the output voltage Vout does not reach a given value. In this step, then, since the circuits of the system on the secondary are not driven properly, they could give rise to energy dissipation as well as to the risk of malfunctioning.