The transmission device in a radio communications system includes a transmission circuit that amplifies a signal obtained from an orthogonal baseband signal and transmits the resultant signal. A transmission circuit with improved efficiency is needed in order to realize greater miniaturization and low power consumption.
Under such circumstances, in the recent years the transmission circuit using a delta-sigma modulator has attracted attention because high power efficiency switching amplifiers, for example those, represented by class-D amplifiers and class-E amplifiers are adopted as power amplifiers. Here in this description, the term ‘delta-sigma modulator’ indicates both the delta-sigma modulator and the sigma-delta modulator.
In general, when orthogonal baseband signals I(t) and Q(t) are subjected to orthogonal modulation with a carrier wave frequency fc from an oscillator by means of an orthogonal modulator, the orthogonally modulated signal e(t) can be represented by the following expression 1.[Math 1]e(t)=I(t)·cos(2πfct)−Q(t)·sin(2πfct)  (Ex. 1)
It is assumed herein that the amplitude (amplitude component) of the baseband signal can be represented by A(t) as in the following expression 2, the phase (angular component) of the baseband signal can be represented by θ(t) as in the following expression 3, the modulated signal e(t) of the above Ex. 1 can be replaced by the following expression 4.
                    [                  Math          ⁢                                          ⁢          2                ]                                                                      A          ⁡                      (            t            )                          =                                                            I                ⁡                                  (                  t                  )                                            2                        +                                          Q                ⁡                                  (                  t                  )                                            2                                                          (                  Ex          .                                          ⁢          2                )                                [                  Math          ⁢                                          ⁢          3                ]                                                                      θ          ⁡                      (            t            )                          =                              tan                          -              1                                ⁢                      {                                          Q                ⁡                                  (                  t                  )                                                            I                ⁡                                  (                  t                  )                                                      }                                              (                  Ex          .                                          ⁢          3                )                                [                  Math          ⁢                                          ⁢          4                ]                                                                                                                e                ⁡                                  (                  t                  )                                            =                            ⁢                                                A                  ⁡                                      (                    t                    )                                                  ·                                  {                                                                                    cos                        ⁡                                                  (                                                      θ                            ⁡                                                          (                              t                              )                                                                                )                                                                    ·                                              cos                        ⁡                                                  (                                                      2                            ⁢                            π                            ⁢                                                                                                                  ⁢                                                          f                              c                                                        ⁢                            t                                                    )                                                                                      -                                                                  sin                        ⁡                                                  (                                                      θ                            ⁡                                                          (                              t                              )                                                                                )                                                                    ·                                              sin                        ⁡                                                  (                                                      2                            ⁢                            π                            ⁢                                                                                                                  ⁢                                                          f                              c                                                        ⁢                            t                                                    )                                                                                                      }                                                                                                        =                            ⁢                                                                    A                    ⁡                                          (                      t                      )                                                        ·                  cos                                ⁢                                  {                                                            2                      ⁢                      π                      ⁢                                                                                          ⁢                                              f                        c                                            ⁢                      t                                        +                                          θ                      ⁡                                              (                        t                        )                                                                              }                                                                                        (                  Ex          .                                          ⁢          4                )            
The above Ex. 4 is equivalent to the result of multiplication between the phase modulated signal P(t) having an amplitude of 1, represented by the following expression 5 and the amplitude component signal A(t) of the above Ex. 2.[Math 5]P(t)=cos {2πfctθ(t)}  (Ex. 5)
Here, when the orthogonal baseband signals I(t) and Q(t) are converted by the following expressions 6 and 7, angular component signals I′(t) and Q′(t) that have an amplitude of 1 and that only show angular component information can be obtained.
                    [                  Math          ⁢                                          ⁢          6                ]                                                                                  I            ′                    ⁡                      (            t            )                          =                  {                                    I              ⁡                              (                t                )                                                                                                          I                    ⁡                                          (                      t                      )                                                        2                                +                                                      Q                    ⁡                                          (                      t                      )                                                        2                                                              }                                    (                  Ex          .                                          ⁢          6                )                                [                  Math          ⁢                                          ⁢          7                ]                                                                                  Q            ′                    ⁡                      (            t            )                          =                  {                                    Q              ⁡                              (                t                )                                                                                                          I                    ⁡                                          (                      t                      )                                                        2                                +                                                      Q                    ⁡                                          (                      t                      )                                                        2                                                              }                                    (                  Ex          .                                          ⁢          7                )            
The phase modulation signal P(t) in Ex. 5 is equal to the signal which is obtained by performing orthogonal modulation on angular component signals I′(t) and Q′(t) with carrier wave frequency fc by means of an orthogonal modulator.
Accordingly, the modulated signal e(t) can be obtained in the following manner. First, orthogonal baseband signals I(t) and Q(t) are separated into amplitude component signal A(t) represented by the above Ex. 2 and angular component signals I′(t) and Q′(t) represented by the above Exs. 6 and 7. Subsequently, angular component signals I′(t) and Q′(t) are orthogonally modulated with the carrier wave frequency fc by means of an orthogonal modulator to obtain phase modulated signal P(t). Thereafter, phase modulated signal P(t) is multiplied by amplitude component signal A(t) to thereby obtain modulated signal e(t).
When, with the above configuration, amplitude component signal A(t) is subjected to delta-sigma modulation as described below, it is possible to realize a high-efficient transmission circuit using a switching amplifier.
FIG. 1 shows a configuration of a related art transmission circuit described in Patent Document 1.
As shown in FIG. 1, the transmission circuit described in Patent Document 1 includes phase modulator 1a, multiplier 2a, delta-sigma modulator 3a, band pass filter (BPF: Band Pass Filter) 4a, pre-processing circuit 6, switching amplifier 7, oscillators 8 and 9. Here, switching amplifier 7 is denoted as “PA (Power Amplifier)” in FIG. 1.
In Patent Document 2, as shown in FIG. 2 a transmission circuit equivalent to the transmission circuit shown in FIG. 1 from which switching circuit 7 is omitted is described. In FIG. 2, frequency modulator 1b, amplitude modulator 2b, delta-sigma modulator 3b, band pass filter 4b and data generator 5 are equivalent to phase modulator 1a, multiplier 2a, delta-sigma modulator 3a, band pass filter 4a and pre-processing circuit 6 shown in FIG. 1.
Therefore, the following description will be given on only the transmission circuit described in Patent Document 1 shown in FIG. 1.
Pre-processing circuit 6 separates orthogonal baseband signals I(t) and Q(t) into amplitude component signal A(t), angular component signal I′(t) and Q′(t) and outputs the resultant signals.
Phase modulator 1a corresponds to an orthogonal modulator and generates phase modulation signal P(t) from angular component signals I′(t) and Q′(t).
Delta-sigma modulator 3a subjects amplitude component signal A(t) to delta-sigma modulation to convert the signal into amplitude pulse modulated signal A′(t).
Multiplier 2a performs amplitude modulation on phase modulated signal P(t) with amplitude pulse modulated signal A′(t). As a result, multiplier 2a outputs a signal obtained by performing on-off modulation on phase-modulated signal P(t) of a fixed amplitude in accordance with information of “1” and “0” of amplitude pulse modulated signal A′(t). Accordingly, the amplitude of the output signal from multiplier 2a takes 0 or a constant value.
Switching amplifier 7 amplifies the output signal from multiplier 2a. The amplitude of the output signal from the multiplier 2a takes 0 or a constant value as stated above. Accordingly, even when the output signal from multiplier 2a is amplified by amplifier 7, the resultant signal will not involve any distortion accompanied by nonlinearity of the power amplifier and the signal can be amplified and transmitted at high power efficiency that switching amplifier 7 can inherently offer.
Here, amplitude pulse modulated signal A′(t) is a delta-sigma modulated signal that is obtained by subjecting amplitude component signal A(t) to delta-sigma modulation through delta-sigma modulator 3a. Therefore, as shown in the following expression 8, amplitude pulse modulated signal A′(t) is given as the sum of the amplitude component signal A(t) as the input signal to delta-sigma modulator 3a and the quantization noise N(t) accompanied by one-bit pulse modulation.[Math 8]A′(t)=A(t)+N(t)  (Ex. 8)
When amplitude pulse modulated signal A′(t) is used to perform amplitude modulation on phase modulated signal P(t) by means of multiplier 2a, the output signal from multiplier 2a also involves quantization noise.
It is assumed that Q(t) represents the quantization noise arising from quantization and Q(z) represents the Z-transform of Q(t). For example, when a first order delta-sigma modulator is employed as delta-sigma modulator 3a, residual noise N(z) in the output of the first order delta-sigma modulator can be represented by the following expression 9 based on the noise shaping characteristics in transmission characteristics of the first order delta-sigma modulator.[Math 9]N(z)=(1−Z−1)·Q(z)  (Ex. 9)
Here, in the above Ex. 9, Z−1 means one sample delay in the clock rate of delta-sigma modulator 3a. 
Accordingly, the Z-transform of the above Ex. 8 is written as the following expression 10.[Math 10]A′(z)=A(z)+(1−Z−1)·Q(z)  (Ex. 10)
In other words, the above Ex. 9 is the expression representing the noise-shaping characteristics of delta-sigma modulator 3a for shifting the noise on the low-frequency side (the transmission signal band) of quantization noise Q(z) that is uniformly distributed over the frequency axis, to the high-frequency side (the outside of the transmission signal band).
Optimization of the noise-shaping characteristics of delta-sigma modulator 3a represented by the above Ex. 9 makes it possible to reduce and to eliminate quantization noise by band pass filter 4a downstream of switching amplifier 7.
Here, the noise-shaping characteristics of delta-sigma modulator 3a is determined depending on the order of delta-sigma modulation and the over sampling ratio (OSR: Over Sampling Ratio). However, if this noise-shaping characteristics have not been optimized, a problem occurs in which noise remaining in the lower frequency (in the transmission signal band) than the stopband of band pass filter 4a spoils the radio characteristics required for the transmission equipment.
Non-Patent Document 1 related to Patent document 1 describes the conditions required for a delta-sigma modulator to yield the characteristics of the desired signal-to-noise ratio for IEEE (Institute of Electrical and Electronic Engineers) 802.11b OFDM (Orthogonal Frequency Division Multiplexing) signals. Specifically, in the example of Non-Patent Document 1, even in the simulation level, a delta-sigma modulator is required to operate at a clock frequency set by an OSR of 32 (i.e., 640 MHz) or by an OSR of 64 (i.e., 1.28 GHz) for the sampling frequency 20 MHz in the original baseband signal.
In general, FPGAs (Field Programmable Gate Array) and ASICs (Application Specific Integrated Circuit) for processing the baseband signal have the advantage in which logic and filter circuits can be easily configured. However, the operation clock frequency (including not only the core but also input and output) of FPGAs and ASICs reach 200 MHz to 400 MHz at the highest even with use of recent miniaturization processes. Therefore, it is impossible with FPGAs and ASICs to realize the desired OSR. Accordingly, the above desired OSR needs to be realized with digital RF circuits that operate at a clock rate of some GHz like the example of Non-Patent Document 2.
Even if an upsampling process and a high-order filtering process that satisfy the aforementioned desired OSR can be realized in a digital RF circuit operating at a clock rate of some GHz, the following problems, still remain.
In the related art shown in FIG. 1, phase-modulated signal P(t) of carrier wave frequency fc, as the output signal from phase modulator 1a, is an analog continuous signal. Further, the carrier wave (frequency fc) from oscillator 9 and the sampling clock (frequency fs) for delta-sigma modulation from oscillator 8 are asynchronous. Even if a common reference generator is used for both signals, when there is duty difference resulting from the difference between the signals (sinusoidal signal and rectangular clock signal) or when the two signals take different paths, it is impossible to make the temporal relationship between the two signals constant. For these reasons, the two signals become asynchronous.
As described above, in multiplier 2a, phase modulated signal P(t) is on-off modulated by amplitude pulse modulated signal A′(t) as the output signal from delta-sigma modulator 3a. In this on-off modulating operation, when the carrier wave (frequency fc) in phase modulated signal P(t) and the sampling clock (frequency fs) for delta-sigma modulation are asynchronous, in order to alleviate its influence, it is necessary to adapt fc to be sufficiently higher than fs.
Also in Non-Patent Document 1 related to Patent Document 1, the condition in which fc should be sufficiently higher than fs (fs<<fc) is specified. However, actually, fs needs to be a frequency ranging from about 600 MHz to about 2 GHz, as stated above. In contrast, the radio frequency fc in UTRA (Universal Terrestrial Radio Access) and E-UTRA (Evolved Universal Terrestrial Radio Access) ranges from 700 MHz to 3.5 GHz. Accordingly, fc forms an asynchronous clock having a frequency approximately equal to or markedly close to fs. In this case, a the problem occurs in which the signal obtained by on-off modulation of phase modulated signal P(t) with amplitude pulse modulated signal A′(t) cannot form a desired pulse waveform, due to timing deviation of signal transitions.
Here, as a technique for synchronizing the carrier wave (frequency fc) in phase modulated signal P(t) with the sampling clock for delta-sigma modulation (frequency fs), phase modulator 1a can be formed of a digital circuit based on a synchronized clock system, instead of an analog circuit. In this case, I′(t) and Q′(t) are upsampled to 4× sampling rate of the carrier wave frequency fc while I′(t) is repeatedly multiplied by 1, 0, −1, 0 . . . and Q′(t) is repeated multiplied by 0, 1, 0, −1 . . . , to thereby realize digital orthogonal modulation. However, fc ranges from 700 MHz to 3.5 GHz as stated above. Accordingly, phase modulator 1a needs to be operated at four times the rate of fc, i.e., 2.8 GHz to 14 GHz. This entails a high level of difficulty and poses many problems.
Patent Document 3 proposes, as a technique for synchronizing the carrier wave in the phase modulation signal with the sampling clock for delta-sigma modulation, the technique illustrated in FIG. 11 of Patent Document 3. Specifically, in this technique, the phase modulated signal that has been orthogonally modulated with the carrier wave by an orthogonal modulator (IQ modulator) is transformed into a pulse phase signal by a pulse phase signal generator, and this pulse phase signal is used as the clock for the delta-sigma modulator. However, in this technique, the sampling clock for delta-sigma modulation has been synchronized with the carrier wave, but cannot have been synchronized with the actual data of the amplitude component signal of the baseband signal input to the delta-sigma modulator. Accordingly, the delta-sigma modulator will perform data transmission between asynchronous clocks. As a result, a problem occurs in which the delta-sigma modulator samples invalid data during the period in which the amplitude component signal, as the input signal, is transitioning.
Here, in the examples of FIGS. 3, 14 and 36 of Patent Document 3, since the amplitude component signal is not a digital baseband signal but is generated from the signal that has been orthogonally modulated by the carrier wave through the orthogonal modulator (IQ modulator), the above problem will not occur. However, since the input signal to the delta-sigma modulator is an analog signal, a 1 bit D/A converter that operates based on carrier wave frequency fc is needed in computing the difference between the output signal from, and input signal to, the delta-sigma modulator in the delta-sigma modulation process, which is difficult to achieve.