The input stage of a digital transceiver typically includes a fully differential programmable gain stage which operates as a low-noise amplifier. The differential programmable gain stage scales a small differential signal present at its input to a full-scale value for analog-to-digital conversion or other processing. The gain (G) of a differential programmable amplification stage is given by:G=R2/R1  (1)where R1 is the resistance between the signal input and the input of the differential amplifier and R2 is the resistance between the input of the differential amplifier and the output. R1 and R2 are often implemented as programmable resistor arrays so that the gain of the amplifier can be adjusted by controlling the number of resistors switched into or out of the input and output resistance networks. For MOS (Metal Oxide Semiconductor) based technologies, R1 and R2 can be made programmable by connecting resistors in series with MOS transistor switches. The MOS transistor switches are in turn connected to the virtual ground nodes of the differential amplifier, i.e., the input nodes of the differential amplifier located between R1 and R2. R1 and R2 can be adjusted by activating or deactivating different ones of the transistor switches.
For many systems, the receiver input is AC-coupled. The common-mode control of the differential amplifier stage senses the average of the single-ended output voltages and forces them to the applied common-mode voltage (VCM). The input nodes of the amplifier are kept at the same potential via R2. However, common-mode sensitivity with respect to linearity is problematic for conventional differential amplifiers. For example, the inputs to the differential amplifier may not only carry the wanted (small) differential receive signal, but also a common-mode signal may appear, e.g., when the external signal ground differs from the on-chip ground, or the receive signal has picked up a common-mode disturbance along the line. Common mode disturbance is present in many types of communication technologies such as power line communication and xDSL, where x is a placeholder for different DSL (Digital Subscriber Loop) technologies.
The differential voltage at the virtual ground nodes of the amplifier stage is very small due to the large differential gain of the amplifier. The situation is different for a common-mode input signal. For operational amplifiers, the common-mode is sensed and controlled at the outputs of the amplifier. As such, the virtual ground nodes can experience a relatively large common-mode excursion, especially at high gain-settings. Assuming an ideal common-mode control at the output of the amplifier, the common mode voltage at the virtual ground nodes of the amplifier is given by:
                              V                      VGND            ,            CM                          =                                            V                              IN                ,                CM                                      ⁢                                          R                ⁢                                                                  ⁢                2                                                              R                  ⁢                                                                          ⁢                  1                                +                                  R                  ⁢                                                                          ⁢                  2                                                              =                                    v                              IN                ,                CM                                      ⁢                          G                              1                +                G                                                                        (        2        )            where νIN,CM is the common mode disturbance at the receiver input. For sufficiently large differential gain G, the common-mode signal at the receiver input is directly transferred to the amplifier inputs. Even at low gain settings, the common-mode signal appearing at the virtual ground nodes of the operational amplifier can be considerably large, e.g., for G=1 (0 dB) a common-mode attenuation of 6 dB results from equation (2).
In a standard implementation, the gates of the MOS transistor switches included in the programmable resistor arrays (R1 and R2) are connected to a positive supply voltage when switched on. Under these conditions, the common-mode voltage present at the source and drain of each activated transistor switch modulates the channel resistance (RON) of the transistor by changing the gate overdrive voltage. For applications requiring high linearity, the linearity of RON is a critical parameter that limits receiver performance. A mixing of the differential input voltage with the common-mode signal occurs at the amplifier inputs when RON is modulated by common mode disturbance, causing reduced receiver dynamic range. The modulation of RON occurs because of signal-dependent fluctuations in the gate-to-source potential of the switch transistor. Such fluctuations in the gate-to-source potential of the switch transistor can arise when the source junction of the transistor is not connected to AC ground and the gate is driven by a DC signal. Modulations in RON cause the total input impedance of the amplifier stage to be modulated by the common-mode input voltage, introducing distortion along the signal path. For example, consider a sine wave common-mode disturber having a frequency of 11.2 MHz and a 30 MHz DMT (Discrete Multi-Tone) differential input signal of interest, representative of a VDSL2-system. As a measure of linearity, the average MTPR (Missing-Tone Power Ratio) at the output of the first receiver stage can be evaluated. MTPR is defined as the ratio of in-band carrier and in-band spurious-tones, eventually generated by nonlinear effects. MTPR is usually measured in the “gaps” of deliberately missing carriers. For an ideal receiver, the average MTPR is above 100 dB in the absence of a common-mode disturber, which is quite adequate for the target system. The linearity quickly drops to problematic levels when a common-mode disturber is applied at the input. Because such common-mode disturbances can appear at random, they may disrupt an established data link, leading to a reduction in the data rate, or even cause synchronization loss.
The effect common-mode disturbance has on linearity can been addressed by providing a common-mode control-loop which acts on the input nodes of the amplifier. However, a common-mode control-loop adds unwanted noise at the sensitive virtual ground nodes of the differential amplifier. Also, the bandwidth of the common-mode control loop limits the maximum frequency of the common-mode disturber that can be effectively suppressed. Large common-mode bandwidths are in principle attainable with considerable overhead in power consumption, causing the circuit to be very noisy.
Other conventional differential amplification stages employ an active control circuit such as a level shifter for controlling the gates of the MOS transistor switches that form part of the programmable resistor arrays. The level shifter forces the gate nodes of the transistor switches to follow the respective source nodes, i.e. the amplifier virtual ground nodes. Although not problematic with the respect to additional noise, the bandwidth of an active control circuit limits the maximum common-mode frequency for which this technique is effective. For example, in low-noise applications with small input resistors, the total switch size can be quite large. This represents a considerable capacitive load for an amplifier that includes an active control circuit such as a level shifter. This limits the range of common-mode frequencies which can be effectively suppressed for a given transistor switch-size and power budget.
For example, again consider a sine wave common-mode disturber having a frequency of 11.2 MHz and a 30 MHz DMT differential input signal of interest. The 30 MHz DMT signal is applied and the average MTPR for an in-band CM-disturber or worst-case MBPR (Missing-Band-Power-Ratio) for an out-of-band CM-disturber can be evaluated at the output of the receiver stage as a measure of linearity. MBPR is defined as the ratio between a representative in-band carrier and out-of-band spurious tones, eventually generated by nonlinear effects (mixing). An active control circuit such as a level shifter effectively compensates for common mode disturbers up to 10 MHz. For common-mode disturber frequencies above 10 MHz, the level shifter performs even worse than if no active gate control is provided, i.e., where the gate nodes of the MOS transistor switches are tied directly to the positive supply VDD. Increasing the bandwidth of the active control circuit is possible, but at the expense of additional power consumption. Common mode disturbers can of course also be filtered externally to the receiver, but this increases the Bill of Material (BOM) and therefore the system cost.