text:
Luca Turilli Live Call-In Today
Luca Turilli will call in live to the show today! Tune in live from 5-7pm Eastern at http://wruw.org for the live stream, and Luca should be on the air by about 6:15. Submit your questions or song requests via Facebook @metalmeltdownshow.
Rhapsody will be in Cleveland tomorrow at the Agora along with Primal Fear. As a huge European power metal fan, I really couldn't ask for a better combination!
I'm proud to say that Luca did his first American interview with me on the Metal Meltdown many, many years ago, and I look forward to catching up with him for the first time in several years.
Edu Falaschi Interview
Part 1 of my Edu Falaschi interview will air tonight's show from 5-7pm Eastern (23:00-01:00 CET), with a live stream available at WRUW-FM 91.1 Cleveland.
Edu is celebrating his 25th year as a professional singer, and this new album, Moonlight, offers acoustic takes on some classic songs, including Angra favorites like 'Nova Era' and 'Heroes of Sand' from Rebirth.
Join me tonight to hear songs from Moonlight and the first half of this look back at Edu's career.
Power Quest Interview (8 Apr)
I'm thrilled to welcome PQ back to the Meltdown! Steve Williams and Ash Edison join me on today's show to discuss the relaunch of the band after a 3-year hiatus. The show will air from 5-7pm Eastern (23:00-01:00 CET) with a live stream at WRUW-FM 91.1 Cleveland.
Ash is the new singer of PQ and is also currently in UK-based Dendera. Steve recently left Eden's Curse, with whom he had toured and recorded the last two albums. As Steve did his last PQ interview with me three years ago, I am honored that he and Ash chose my show for their first interview with the new lineup.
It is always a pleasure to have Steve on the show, and I hope you'll join me this evening to hear this new interview.
Sabaton In-Studio (April 11, 2014)
On today's Metal Meltdown from 5-7pm Eastern and 23:00-01:00 CET, Dr. Metal will be joined in-studio by members of Sabaton. They'll join me for the middle of the show, and we'll talk about their current North American tour, forthcoming release, Heroes, and everything else.
Also happening today is the annual telethon show where we'll raise money for WRUW-FM 91.1 Cleveland, the station which has hosted the Metal Meltdown since 1999. It is a non-commercial, non-profit station which relies on listener support for half of its operating budget. We have some great thank-you gifts, so tune in to the show and keep an eye on our Facebook page - @metalmeltdownshow for more information.
The show will be on live at WRUW.ORG today.
Today's Show (April 4, 2014
Big show for you guys today! Join me at wruw.org from 5-7pm Eastern, 23:00-01:00 CET.
Piet Sielck of Iron Savior (first time in over 10 years)
Ricky Warwick of Black Star Riders
Dave Silver of Savage Messiah
You'll also hear new music from Sonata Arctica and Gamma Ray, among others. Check in via Facebook (metalmeltdownshow) during the show or at 216-368-2208.
Also, next week is Telethon, WRUW-FM's annual fundraiser during which we ask for financial support from our listeners to help keep this noncommercial, non-corporate radio station thriving. Not only can you support a good cause, but you can pick up some great premiums as thank-you gifts. Keep watching this page for more info, but here's a small list of some of the items available:
Tarja: Act 1 DVD
Gamma Ray: Skeletons & Majesties Live DVD
Volbeat: Outlaw Gentlemen and Shady Ladies (2013 Album of the Year)
Power Quest: bootleg DVD of their final show ever
Sonata Arctica: autographed single 'I Have a Right'
Today's Show (March 28, 2014)
Heri Joensen of TÝR will appear on today's Metal Meltdown on WRUW-FM 91.1 Cleveland (live @ wruw.org) from 5-7pm Eastern. I sat down backstage at the House of Blues on Sunday night when Heri and the band were opening up for Death Angel and Children of Bodom.
In addition, you'll hear from Sacred Blood, a Greek metal band whose lyrics are based completely in Greek history and literature. Their 2012 release, Alexandros, focuses entirely on the life of Alexander the Great, and their 2014 effort, Argonautica, will be about Jason and the Argonauts. I caught up via Skype with Polydeykis, the guitarist, and Marios, the band's lyricist.
Finally, I'll have new music from Sonata Arctica, Gamma Ray, the Dio tribute album, Delain, Axxis, and many more.
Send your song requests to me via Facebook (metalmeltdownshow) or at 216-368-2208.
I'll be interviewing Chris Bay of Freedom Call on today's Metal Meltdown with Dr. Metal. Tune in at wruw.org from 22:00-midnight CET (5-7pm Eastern time). The band's new album, Beyond, is out right now on SPV and is really one of their best so far with many catchy songs.
Get your song requests in to me at Facebook.
New music from Elvenking, Sonata Arctica, Gun Barrel, Lacuna Coil, Gus G., and many more.
Today's Show (November 29, 2013)
Two big interviews on today's Metal Meltdown - tune in live at wruw.org from 5-7pm Eastern. First up is a conversation with David Ellefson of MEGADETH, who just released his autobiography, My Life with Deth. Then you'll hear from Gareth Murdock of ALESTORM, who came through Cleveland this week on their first headlining tour of the US. Song requests for either band (or anything else) can be submitted at the Facebook page.
Gareth and Dr. Metal at WRUW
Steve Williams (ex-Power Quest) and Paul Logue of Eden's Curse will join me on today's Metal Meltdown from 5-7pm Eastern (23:00-01:00 CET). Tune in live at wruw.org or WRUW-FM 91.1 Cleveland to hear this interview and several songs from the new album, Symphony of Sin, now available on AFM Records. Submit your requests for today's show on Facebook.
Interview with JOE COMEAU (Duskmachine/Liege Lord)
After stints in Overkill and Annihilator working on some great albums, Joe Comeau has joined forces with Duskmachine, lending his vocals for their new self-titled release. Liege Lord has reformed, too, with plans to record their first album since 1988's Master Control. Joe and I talked about both bands and a little bit about his history in metal. It was also very interesting to hear about his day job and how it relates to his music, as you don't often hear musicians talk about what they do to pay the bills. There's some good stuff in here.
Check out the full audio or text highlights below.


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The Blue Berry Eyebright™ tablets contain blueberry, etageres flower standardized to contain 10 mg lutein per daily dose, and eyebright. Lutein is a carotenoid naturally found in the macula of the eye - the part of the eye responsible for central vision.


text:
What is Pavilion2022?
Budding Konta's and Murray's
Quite simply Pavilion2022 is the project initiated and led by Fernhurst Recreation Trust with the support of the football, stoolball , tennis and cricket clubs to replace the existing pavilion with one that meets the demands the sports governing bodies and future generations.
The Last Decade
Fernhurst has transformed its recreation ground and participation in sport in the last decade. Where once only a handful of children played on the Rec, now 300 are regularly involved with football, cricket and tennis. After 50 years of service and with almost 800 members the pavilion is creaking at the seams. This vital sporting hub for our rural community has outgrown the facilities and we want a pavilion that can engage even more local residents in sport for the next 50 years.
Current pavilion
A brand new, well-drained football pitch along with cricket nets and tennis courts have all been installed thanks to charities, councils, sports bodies and villagers. The final piece in the puzzle is the pavilion. It was opened in 1971 and twice extended in the 1980's. We aim to both celebrate and decommission it on its 50th anniversary and to open its replacement in 2022.
The work of the Trust and the sports clubs has already achieved 'the numbers' in participation. Subscriptions are among the lowest around ensuring a true 'sport for all' philosophy.
Our strategic importance is clear from the number of families not just from the village but also those who travel in from Haslemere, Lodsworth, Midhurst, Petworth, Camelsdale, Grayshott, Hindhead, etc. In total, 792 members are registered to play sport in Fernhurst. We are already punching well above our weight with bigger clubs as Fernhurst children have been selected for district and county teams.
Recruit, retain, modernise
The task now is clear: the tired and outdated facilities have to be replaced simply to serve and to retain the people we already have. The existing pavilion was built for another age and for far fewer people. It has been patched up to comply with modern regulations on disabled access, food hygiene, child welfare and the like. The public conveniences could and should be much better.
The aim is to raise £1 million to build a pavilion which complements our beautiful setting. Fernhurst Recreation Ground Trust and the affiliated sports clubs have already raised tens of thousands and are committed to more.
Early drawing
Local architects have drawn up designs which greatly improve the communal areas, function rooms, kitchen, bar and viewing gallery. Changing rooms will be expanded with separate areas for juniors. The public conveniences will also be distinct from those for sports teams. It will be a genuine 'multi use' building allowing different clubs to hold simultaneous events.
Fernhurst's people have already demonstrated their passion and commitment. A village where children didn't have much to do has been transformed to one which now attracts families from outside. With a new pavilion Fernhurst sports clubs can secure their position as one of the flagship multi-sports facilities in the area. We have the volunteers and the vision. We invite you to join us in this final part of the transformation so Fernhurst can have a pavilion to be proud of.
Pop over to the CONTACT page to leave a message of support.

redpajama_set_name: RedPajamaCommonCrawl


text:
Qube set to build 162-room 'modular' hotel in Perth
Accommodation operator Qube is set to build a 162-room 'modular' hotel near Perth's domestic airport with an investment of $20m, in an effort to offer cheap accommodation targeted at the fly-in fly-out market of Western Australia.
By Shalini Nair
Accommodation operator Qube is set to build a 162-room 'modular' hotel near Perth's domestic airport with an investment of $20m, in an effort to offer cheap accommodation targeted at the fly-in fly-out market of Western Australia.
Claimed to be the first hotel of its kind in the country, the modules will be constructed in China and then shipped to the city.
The project will be located at Ascot, approximately 1.2km from the airport.
Construction of the first phase of the project is expected to be completed in three months and is likely to be opened by June 2013, reported The West Australian.
"The modules will be constructed in China and then shipped to the city."
Qube sales general manager Lynne Jackson was quoted by the publication as saying that modular hotels could transform the hospitality industry as they can be built in just three months.
Modular construction to build hotels and offices is growing increasingly popular in Asia and Europe as it is believed to be cost-effective and quicker; in contrast, it takes up to three years to build a hotel in the traditional way.
The new hotel will also mark the company's long-term plans to expand into the mining regions in the northern part of Western Australia to capitalise on acute accommodation shortages.
The hotel will feature an eight-hour rate option in order to provide flexibility to mining firms.
Meanwhile, Qube's separate project near the airport, Aurum Hotel & Residences featuring 52 serviced apartments, is set to open in March 2013.
Perth has the highest hotel occupancy in the country, with an average rate of 86%.
Many new hotel projects are underway in the city such as a $16m serviced apartment complex, and a 300-room hotel that will be operated by international hotel group Starwood Hotels and Resorts.
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Blackstone explores sale of G6 Hospitality
Wyndham Hotels & Resorts adds The Bristol Panama to Registry Collection Hotel
IHG Hotels & Resorts plans to open 39 hotels in India
Increase in high-net-worth individuals bodes well for luxury tourism


Home Design 900 Square Feet #0 - 900 Sq Ft Home Floor Plans was uploaded in July 19, 2018 at 10:58 am. If you want to apply this picture as your computer background, you can click the download link beneath or you can just right click on the image above , then select "Save Image As" to save the Home Design 900 Square Feet #0 - 900 Sq Ft Home Floor Plans or by choose the "Set Desktop Background As" choice if your internet browser has this option. If you were no able to bring the the best image you are looking for, you should go for "Search Column" at top right or browse other photo backgrounds whatever you wish. This picture has the picture size is 68kB and the dimension of 575x570. Home Design 900 Square Feet #0 - 900 Sq Ft Home Floor Plans has been seen by 71 users and it is an attachment from Home Design 900 Square Feet.


A knowledge paper designed to challenge you and help you evolve spiritually.
These are specially handpicked books and papers from thousands of sources that are tailored to help one grow substantially.
A power spiritual tablet that will help you evolve your consciousness and change your life in the areas of prosperity, health, protection, relationships, and more!
A special bonus item – it could be anything from a divine letter to a specially crafted video for entertainment/evolution.
Access to the Forum Focus area only visible to members of this site.
Here you can ask any spiritual question they you have and it will be discussed among peers who have similar desires to learn and grow. Take advantage of a judgment free zone to ask questions you may not feel comfortable asking anywhere else.
Facebook Updates of uplifting posts and new releases.
A calendar of the upcoming events.
We have been gifted with special work from the Order of the Red Dragon.
This is Initiate level work.
It is unpublished information that can only be received from The Order.
This information can only be accessed by members of Tybro Knowledge Papers and should NOT be shared.
We will begin releasing these as Knowledge Papers for Level 7 members and up, starting June 2016.


text:
\section{Introduction}\label{sec:intro}

The second generation of gravitational-wave detectors Advanced LIGO~\cite{aLIGO2015} and Advanced VIRGO~\cite{Acernese_2014} have been observing the compact binary mergers for over five years. The first direct detection of gravitational waves from the inspiral and merger of two binary black-holes happened on September 14, 2015, opening up the field of gravitational wave astronomy~\cite{GW150914}. Since then there have been two sets of upgrades to improve the sensitivity of the Advanced LIGO and VIRGO detectors~\cite{comm_O3} and the three observational science runs have confirmed over 50 gravitational wave signals from binary collisions~\cite{abbott2019gwtc, abbott2020gwtc}.

The Advanced LIGO and VIRGO detectors are dual-recycled Fabry-Perot Michelson interferometers capable of measuring relative peak displacement between the two interferometer arms to less than one attometer~\cite{aLIGO2015, Acernese_2014}. To enable this precise measurement, the core and auxiliary optics are seismically isolated and the feedback loops are set up to actuate on translational and angular degrees of freedom~\cite{aLIGO2015}. The error signals to control the optics are derived using Pound-Drever-Hall (PDH) technique~\cite{black2001introduction, fritschel1998alignment} from 9~MHz and 45~MHz sidebands modulating the carrier laser beam.
These error signals are acquired using radio-frequency (RF) photodiodes - single-segment diodes for length sensing, and four-segment quadrant photodiodes for angular sensing. Similarly, beam waist position and size sensing can be done using bulls eye segment diodes \cite{mueller2000determination}, mode converters \cite{magana2019sensing} and aperture-based schemes \cite{miller2014length}. Alternatives to this readout scheme using jitter modulation techniques have also been proposed \cite{Fulda:17}.

Since any degradation in the optical wave front will degrade these signals, significant effort has gone into ways to image the RF optical beat pattern of the readout beam for diagnostic purposes. Previously developed sensors for the LIGO and Virgo detectors include scanning-type \cite{Goda:04, van2016advanced, agatsuma2019high} and optical lock-in \cite{Cao:20, brown2021differential} phase cameras, all of which require additional optical elements ahead of the photo sensor.

In our work, we redesign the traditional time-of-flight camera to directly record these amplitude-modulated laser beat signals in QVGA resolution (320 $\times$ 240 pixels). We achieve this by phase locking the OPT8241 time-of-flight camera to the external reference oscillator that modulates the laser.  This approach has the benefit of requiring minimal optical components and beam shaping,
making the CMOS phase camera an excellent diagnostic tool for gravitational-wave detectors that can be easily deployed at any optical port of the interferometer. We also demonstrate that this phase camera can be used for generic wavefront sensing.

The sensitivity of interferometric gravitational wave detectors such as Advanced LIGO scales with the amount of power circulating in the arm cavities. Following the O3 upgrade, the Advanced LIGO detectors reached 200~kW of power in the arm cavities~\cite{comm_O3} for the first time. The high power in the arm cavities exposes point defects in the coatings of the test masses. Under high power these point defects, or point absorbers, burn into the coating of the Advanced LIGO test masses~\cite{brooks2021point}, locally heating and deforming the optic. This has proven to be problematic, limiting power buildup in the arm cavities and dark port contrast, both of which limits the sensitivity of the Advanced LIGO detectors. Specifically, the thermo-refractive and thermo-elastic surface deformations due to the point absorbers induce optical path distortions affecting the phasing of the carrier and the modulating sidebands~\cite{brooks2021point}. For the carrier, this detuning results in higher-order spatial modes being resonant in the arm cavity, causing a loss in the detector sensitivity. Additionally, the non-resonant RF sidebands couple into higher-order spatial modes and degrade the alignment and control error signals error.

We aim to use the CMOS phase camera to image the deformations in the laser beat signals that arise from these point defects. In the paper, we briefly describe the principle of operation of time-of-flight cameras in section~\S\ref{sec:ppl_ToF}. Next, in section~\S\ref{sec:electronics} we describe the hardware and software changes made to reconfigure the time-of-flight camera into a CMOS phase camera. The experimental layout to test the sensitivity of the phase camera and the corresponding noise model to measure amplitude modulated signals is described in section~\S\ref{sec:exp_sense}. We model the point absorber to estimate the change in beat signal and correlate it with the sensitivity of the camera in section~\S\ref{sec:pointAbs}. Lastly, we discuss the applicability of the CMOS phase camera in Advanced LIGO and A+ detectors in section~\S\ref{sec:results}.

\section{Principle of Operation of time-of-flight cameras}\label{sec:ppl_ToF}

\begin{figure}[htp]%
\center
\includegraphics[width=0.9\linewidth, frame]{PC_fronthigh.jpg}
\caption{The prototype CMOS phase camera based on the commercially-available OPT8241 time-of-flight sensor. The custom-built enclosure ensures shielding from interference from unwanted RF signals.}
\label{fig:pcdemo}
\end{figure}

Time-of-flight cameras employ a 2D array of pixels with depth sensing capabilities to independently measure the distance of a particular object in the field of view. Commercially available time-of-flight cameras typically have integrated illumination sources. They rely on either pulse-modulation, directly measuring the pulse travel time, or on continuous-wave amplitude-modulation, measuring a phase delay via demodulation with a local reference. The measured phase delay provides depth information. Since we intend to use the sensor to record the beat map of laser fields we selected the second type sensor for our research.

The amplitude $A$, phase $\phi$ and DC value $I_0$ of the amplitude modulated beam can be found by measuring the gated photo current $C$ at four demodulation phases ($\Phi$ =  0, $\pi/2$ , $\pi$ , $3\pi/2$)~\cite{lange:00}. We have
\begin{equation}\label{eqn:amp_rf}
    \text{A}= \frac{\sqrt{\{C(0) - C(\pi)\}^2 + \{C(\pi/2) - C(3\pi/2)\}^2 }}{2}
\end{equation}
\begin{equation}\label{eqn:phase}
    \phi= \measuredangle [(C(0) - C(\pi))~+i(C(\pi/2) - C(3\pi/2))]
\end{equation}
\begin{equation}\label{eqn:amp_dc}
    I_0=\frac{C(0) + C(\pi/2) + C(\pi) + C(3\pi/2)}{4}
\end{equation}

\section{Motivation and Design}\label{sec:electronics}

For application as a wavefront sensor in gravitational-wave interferometers the time-of-flight camera needs to accept the interferometer RF local oscillator signal as an external reference oscillator source for demodulation.
In continuous-wave amplitude-modulated time-of-flight cameras, both the illumination source and pixels are typically driven using the same internal, radio-frequency (RF) oscillator source. \citet{Shrestha:2016} show that the time-of-flight camera OPT8221 offers flexibility for operation as a standalone demodulation camera with an external signal driving the illumination source and the demodulation within the camera.

The design proposed here uses the OPT8221 camera evaluation board (OPT8241-CDK-EVM), which consists of two primary components: the sensor (OPT8241) and a programmable controller (OPT9221). The OPT9221 is a companion chip to the OPT8241 and is responsible for setting register functions and processing raw data. The sensor is a standard 320 $\times$ 240 pixel array (QVGA format) capable of operating at frame rates up to 150 frames per second (fps), although live-streaming with Voxelviewer software~\cite{voxelviewer} is limited to 60 fps. Neighboring pixels are separated by 15~$\mu$m and are capable of demodulation up to 100 MHz.
In each pixel the charge carriers are sorted into two separate charge storage wells, depending on the state of the local oscillator \cite{opt8241sysdesignguide,schwarte1997new}. The two wells are simultaneously read out and their difference is digitized, removing the DC component from the RF readout. Charge separation becomes less efficient at higher frequencies \cite{LangeThesis}, leading to a reduction in the demodulation amplitude at higher demodulation frequencies.

The camera board can be programmed to provide 12-bit amplitude and phase, as well as 4-bit ambient values in QVGA format. The OPT8241 also features a 850 nm NIR band-pass filter covering, which has a small transmission of about 5\% at 1064 nm for normal incidence~\cite{opt8241datasheet}.  We use the sensor at 1064 nm, the laser wavelength used in the LIGO, VIRGO, and KAGRA gravitational wave detectors~\cite{aLIGO2015, Acernese_2014, aso2013interferometer}.

For application as an active wavefront sensor the camera board must be reconfigured. First, the internal modulation block of the camera board is disabled via the appropriate configuration of register settings in the OPT9221, allowing it to accept an external reference signal. This reference signal needs to be appropriately stepped in quadrature phase, with $\Phi = 0^\text{o}, 90^\text{o}, 180^\text{o}, 270^\text{o}$. The frequency of the reference signal can also be dynamically changed between video frames, permitting imaging beat signals from multiple sidebands simultaneously.
Thus, the CMOS phase camera is capable of measuring both the spatial and temporal characteristics of the illuminating beam in real-time.

\begin{figure}
    \centering
    \includegraphics[width=\linewidth]{Wiring_Diagram.pdf}
    \caption{An overview of the phase camera signal chain and design components. The microcontroller acts as the host device controlling waveform generation and image frame capture. The microcontroller initializes the direct digital synthesizer board. Quadrature phase-stepped signals are generated on the four analog output channels of the direct digital synthesizer board. These signals are sent to an RF switch, which passes the appropriate signal via CMOS converter and gating switch to the camera. This gating switch removes the RF signal during the sensor readout to reduce the electronics noise.
    Every frame is initiated by a frame initialization trigger sent by the microcontroller to the camera board. The camera responds by sending a series of quadrature pulses back to the microcontroller, which are used in an interrupt sequence to set the control signals for the RF switch. }
    \label{fig:signalchain}
\end{figure}

In this section, we present the schematic design for the CMOS phase camera. The various electronic components, along with their functionality, are listed below and illustrated in Fig.~\ref{fig:signalchain}:
\begin{itemize}
    \item {\it Microcontroller}: We use the ARM Cortex-M4-based STM32F407 as a master device. It is programmed to control waveform generation on the Direct Digital Synthesis (DDS) board. The microcontroller transmits a data sequence to initialize each of the individual channels on the DDS board via SPI protocol.
    The microcontroller also initiates the frame capture sequence by sending an initial trigger pulse to the camera board. Depending on the register settings of the OPT9221 controller, the camera board responds by sending a series of quadrature (quad) exposure pulses, which trigger an interrupt sequence on the microcontroller. This interrupt sequence is used to set the logic levels on the RF switch and/or the synchronization flip-flop to output the appropriately phase-stepped signal corresponding to the quad exposure pulse.

    \item {\it Direct Digital Synthesizer (DDS) board}: Waveform generation is handled by the Analog Devices Direct Digital Synthesizer board (AD9959). The AD9959 has four independent  synchronized output channels with tunable amplitude, frequency and phase. The external reference local oscillator used to modulate the interferometer input laser serves as the reference clock signal to the DDS board for synchronization.
    The microcontroller sets frequency, phase and amplitude registers on the DDS board for each channel and activates the settings via a separate I/O-update line.
    For normal operation the four channels are set to a phase of $\Phi =0^\text{o},~90^\text{o},~180^\text{o},~270^\text{o}$ with respect to the reference clock at the desired demodulation frequency during initialization.

    \item {\it RF switch}: A single-pole, four throw RF switch from Mini-Circuits (JSW4-272DR+) is used to switch between the four phased channels from the DDS board, selecting the camera demodulation phase. The RF switch is controlled by the microcontroller. The DDS board and RF switch together can provide an externally referenced and phase-stepped oscillator between 5~MHz to 500~MHz.

    \item {\it Synchronization Flip-Flop}: The DDS board register update is only synchronized to the internal reference clock (SYNC\_CLK), which is running at a higher frequency than the external reference signal. To avoid random phase jumps on update, a flip-flop is used to trigger the update of the DDS registers synchronized with the rising edge of the external reference signal. There are two schemes to operate the phase camera.
    First, the four channels of the DDS board are set up during camera initialization, and the flip-flop synchronization to the reference oscillator guarantees the same relative phase on every camera startup. The RF switch is then used to select the channel with the desired demodulation phase. Since this method does not rely on a synchronized register update during operations it is more robust, but also limited to four demodulation phases and a single frequency.
    In the second approach, we use the flip-flop to actively update the registers and phase-step the output of one of the DDS channels in accordance with the quad pulses from the camera. This scheme, discussed in-depth in section~\S\ref{appsec:ff_scheme}, allows the camera to image the amplitude-modulated beat at different frequencies in real-time, and permits camera operation with six quads, reducing potential cross-talk from harmonics of the modulation frequencies.

    \item {\it CMOS Converter}: To convert the sinusoidal reference signal to a CMOS logic level clock, i.e.~`squaring the clock', we use a high-speed clock distribution integrated circuit (Analog Devices AD9513).

    \item {\it Gating Switch}: We use an additional high-speed buffer chip (Texas Instrument's SN74LVC126A) in the signal chain to gate the output demodulation waveform during each quad exposure. Turning the RF signal off during the readout phase between camera exposures significantly reduces the camera electronics noise.
\end{itemize}
All firmware was developed within the Keil $\mu$Vision IDE and flashed to the microcontroller via a USB interface. This allows the microcontroller to run as the master device after an initial reset event.

\section{Performance Characterization}

\subsection{Experimental Setup}\label{sec:exp_sense}

\begin{figure}
\center
\includegraphics[width=\linewidth]{PCammodsetup.pdf}
\caption{The experimental layout used for testing and quantifying the noise levels of the phase camera. The laser is stabilized by locking to the pre-mode cleaner cavity using the PDH technique. The amplitude-modulated beat signal is generated in the reflection of the cavity when it is locked to one of the 25~MHz sidebands. The bbPD1 is the photodiode sensor used for PDH locking. The bbPD2 is used in calibration and measures the power modulation index of the beam incident on the camera. The total power incident on the camera is attenuated to $\sim$10$~\mu$W.}
\label{fig:exp_setup}
\end{figure}

To characterize the performance of the CMOS phase camera we used the experimental setup shown in Fig~\ref{fig:exp_setup}.
A reference triangular cavity, previously used as
pre-mode cleaner cavity (PMC) in the initial LIGO interferometers until 2010, is used to stabilize the frequency of a 1064 nm Nd:YAG laser via Pound-Drever-Hall (PDH) locking. The cavity has a finesse of 165 and features a piezoelectric-actuated mirror at the apex which allows for tuning of the cavity resonance. The carrier field, $E_{c}=E_{0}e^{i\omega_{c}t}$, is phase-modulated using a resonant electro-optic modulator referenced to a 25-MHz local oscillator.  This adds sidebands to the optical carrier field offset at the modulation frequency $\Omega/2\pi$. The peak-to-peak drive voltage of the local oscillator determines the amplitude of the phase modulation index $\Gamma$. The field incident on the cavity is given by
\begin{equation}
    E_{in} = E_{c}e^{i\Gamma \cos\Omega t} \approx E_{c} \left(1 + \frac{i \Gamma}{2} e^{i \Omega t} + \frac{i \Gamma}{2} e^{-i \Omega t} \right)
\end{equation}
To produce a significant amount of amplitude beat signal between carrier and sideband the cavity length was tuned to resonate on a single sideband. Under the simplifying assumption that one sideband passes the cavity $100\%$, while the other light gets reflected $100\%$, the reflected field becomes to first order
\begin{equation}
    E_{ref} \approx E_{c} \left(1  + \frac{i \Gamma}{2} e^{-i \Omega t} \right) ,
\end{equation}
which is then passed through a 50/50 beam splitter where half of the laser light is reflected onto a single-element broadband RF phototdiode (bbPD1) for PDH locking and the other half is directed to the camera. From here, the beam power is split again with 90\% of the light going to the camera and 10\% incident on a second photodiode (bbPD2) used for calibration -- discussed more in section~\S\ref{subsec:calibration}.  The power incident on the camera is thus amplitude modulated with power modulation index $\Gamma$:
\begin{align}\label{eqn:powermodulation}
    \nonumber
    I_{PC} &= |E_{ref}|^{2} \\
           &\approx |E_{c}|^{2}\big(1 + \Gamma_{} \sin\Omega t\big)
\end{align}
Frequency terms greater than the modulation frequency are filtered by the bandwidth of the camera.
In this single-sideband scenario power modulation index $\Gamma$ is also related to the sideband-to-carrier ratio ${\rm SCR}$ via
\begin{equation}\label{eqn:dBc}
    {\rm SCR \, [dBc]} = 20 \log_{10} \left( \frac{\Gamma}{2} \right),
\end{equation}
where we use the IEEE's definition \cite{807679}.
Unlike typical photodetectors, the measured photocurrent is demodulated at $\Omega$ in each pixel on the sensor array.
The corresponding amplitude and phase maps are then constructed using the measured values of I and Q for each frame. The CMOS phase camera in our experiment was set to capture at a frame rate of 7~Hz with total exposure time of 32~ms, although frame rates of up to 60~Hz are supported.

\subsection{Calibration}\label{subsec:calibration}

\begin{figure
\center
\includegraphics[width=\linewidth]{DCdata_wfit.pdf}
\caption{The DC intensity profile, which is used in calibration, is plotted as a function of sensor position. The residual of the Gaussian fit to the DC profile is shown in the plane on the bottom. The dominant 4-bit digitization noise in the DC output is noticeable. In contrast, the RF output is digitized with 12-bit resolution.}
\label{fig:dcfit}
\end{figure}

For each frame of the 320 $\times$ 240-pixel array, the respective amplitude, phase, and ambient readout channels of the camera are expressed in terms of counts of the analog-to-digital converter or digital numbers (DN). We perform a calibration procedure to determine a calibration factor, $\kappa$, which relates the total number of generated photoelectrons in an image to the corresponding DN value. The calibration procedure is as follows:
\begin{enumerate}
    \item We use the broadband photodiode (bbPD2, see Fig.~\ref{fig:exp_setup}), to determine the modulation index of the beam incident on the camera. We measured its DC transimpedance $Z_{dc}$ to be 1975~$\Omega$ and its RF transimpedance $Z_{rf}$ at 25~MHz, the frequency of the local oscillator, to be 4750~$\Omega$. Then, the power modulation index ($\Gamma_{m}$) of an arbitrary beam incident on \textit{bbPD2} can be calculated via
    \begin{equation}\label{eq:mod_index}
        \Gamma_m = \frac{1}{2} \cdot \frac{V_{pp}^{rf} \cdot Z_{dc}}{ Z_{rf} \cdot V_{dc} }
    \end{equation}
where the peak-to-peak RF signal ($V_{pp}^{rf}$) and the mean DC signal ($V_{dc}$) measured by \textit{bbPD2} are read on an oscilloscope. The factor of 1/2 converts peak-to-peak to amplitude.
    \item To quantify the number of photons incident on the camera we measure the incident power ($P_{in}$) using a calibrated Thorlabs power meter. Since the NIR filter on the camera has a 5$\%$ transmission at 1064~nm at normal incidence~\cite{opt8241datasheet}, the total power incident on the camera after the filter ($P_{in}^{sensor}$) can be calculated. The ambient channel of the camera measures the profile of the incident beam. We fit a Gaussian profile normalized to the $P_{in}^{sensor}$ to get the Gaussian beam parameters and calculate the power density of the beam incident on the sensor, $p_{in}^{sensor}(x,y)$, see Fig.~\ref{fig:dcfit}. The number of photons per area incident on the sensor during the exposure time $T_{exp}$ can then be estimated by
    \begin{equation}\label{eq:no_photons}
        n_{p} (x,y) = \frac{p_{in}^{sensor} (x,y) \cdot T_{exp}}{h\nu}
    \end{equation}
    The exposure time is the sum of each of the quadrature exposure time during which the signal is integrated.
    \item The number of photoelectrons generated in each pixel is related to the number of photons via the quantum efficiency $\eta$, which for the OPT8241 sensor is approximately 2$\%$ for 1064 nm light ~\cite{LangeThesis}. Thus we have
    \begin{equation}\label{eq:no_electrons}
        N_e = \int_{A_{pixel}} \!\!\!\!\!\!\!\!\!\!\!\! n_{e} (x,y)~dx dy = \int_{A_{pixel}} \!\!\!\!\!\!\!\!\!\!\!\! \eta \cdot n_{p} (x,y)~dx dy
    \end{equation}
    \item Finally, the calibration factor $\kappa$ is given by the ratio of the total number of photoelectrons generated to the sum of digital numbers reported in a region of interest (ROI):
    \begin{equation}\label{eq:calib}
        \kappa_{DC} (e^{-}/DN) = \frac{ \Sigma_{ROI} (N_{e}) } {\Sigma_{ROI} (DN_{DC})}
    \end{equation}
    Similarly, the AC calibration is given by
    \begin{equation}\label{eq:calib}
        \kappa_{AC} (e^{-}/DN) = \Gamma_m \frac{ \Sigma_{ROI} (N_{e}) } {\Sigma_{ROI} (DN_{AC})},
    \end{equation}
    where $\Gamma_m$ is the modulation index from Eq.~\ref{eq:mod_index} and we divide by the camera output in the AC readout, $DN_{AC}$.
    We estimate a calibration factor of $\kappa_{DC}$ of $1.9\cdot10^5~e^{-}/DN$  and $\kappa_{AC}$ of $1.0\cdot10^{3}~e^{-}/DN$.

\end{enumerate}

\subsection{Noise Sources}\label{subsec:noise}
The CMOS phase camera offers both the AC and DC readout channels, which are susceptible to noise sources typical to CMOS image sensors. These noise sources can either be temporal or spatial. Temporal noise arises due to the electronic noise from resistive components (i.e. pixel reset noise), dark noise, photon shot noise and other electronics noise related to the image sensor. Spatial noise is primarily due to the pixel-to-pixel imperfections, that is gain and threshold variations in the sensor array.

For the DC readout, the 4-bit digitization of the ambient channel results in large analog-to-digital rounding errors. As a result, the ADC quantization noise is the limiting source of noise for the DC readout. In contrast, the AC readout provides a 12-bit resolution, sufficient for resolving small fluctuations of the measured signal. For the AC readout, the different noise sources of the phase camera are presented below:

\begin{enumerate}
    \item \textit{Shot Noise} in the camera translates to an effective fluctuation in the measured number of photoelectrons generated in the AC readout channel.
    The shot noise scales as the square root of the number of electrons generated during the exposure time. The AC shot noise can be estimated as
    \begin{equation}
        \sigma_{SN}~(e^-) \approx \sqrt{2 N_e}
    \end{equation}
    The factor of $\sqrt{2}$ is due to the demodulation. Shot noise is a fundamental limitation, but our CMOS phase camera is not shot noise limited.
    \item \textit{Electronic Noise} includes all noise sources involved in charge conversion and signal processing within the image sensor. Two prominent noise sources in the CMOS-based sensors are the amplifier noise and the pixel reset noise. Both arise due to the Johnson noise associated with the reset transistor in each pixel. Therefore, the electronic noise (in DN) can be estimated from non-illuminated regions in the frames, using the temporal standard deviation across multiple frames.
    The effective electronic noise can be reduced by frame- and/or pixel-averaging (i.e. temporal or spatial averaging).
    \item \textit{Fixed Pattern Noise} (FPN) describes the spatial noise associated with non-homogeneity between neighboring pixels on the sensor. Fixed pattern noise is dependent on the signal on the sensor and for a given signal can be expressed as~\cite{georgiev2015fixed}
    \begin{equation}
        \sigma_{FPN} = \alpha~\Tilde{C} + \beta
    \end{equation}
    where $\alpha$ is the gain and $\beta$ is a column or row offset in the readout. $\Tilde{C}$ represents the true DC signal with additive Gaussian noise, which represents the temporal fluctuations of the signal contributing to the fixed pattern noise due to the gain. Under uniform illumination, one can estimate the pixel fixed pattern noise (in DN) as the spatial standard deviation from the mean. Using the calibration factor calculated in Section~\S\ref{subsec:calibration}, we can express the fixed pattern noise in terms of the number of electrons. As $\sigma_{FPN}$ varies as a function of the signal intensity, trivial background frame subtraction will not work to improve the SNR. Spatial averaging of frames can reduce the noise at the cost of signal resolution. Technique to subtract fixed pattern noise from phase and amplitude readouts is developed by \cite{georgiev2015fixed}.

\end{enumerate}
Ideally, the limiting noise sources for the CMOS phase camera have a Gaussian distribution~\cite{toronov2003optimization, rapp2008theoretical}. Under this assumption, the variance of each of the quadrature measurements is constant ($\sigma^{2}$). In this case, the total noise in the amplitude ($\sigma_{A}$) can be estimated as the quadrature sum over individual noise sources. The total phase noise ($\sigma_{\phi}$) is estimated as the ratio of the total noise in amplitude to the mean amplitude of the signal, as phase and amplitude noise should be uncorrelated.

\begin{comment}
\begin{figure*}[!ht]
    \centering
    \includegraphics[width=0.49\textwidth]{acimage.pdf}
    \includegraphics[width=0.49\textwidth]{acimagesub.pdf}
     \caption{(a) Reconstructed amplitude of the heterodyne beat created between the TEM00 carrier and reflected sideband field. (b) Residual amplitude noise after subtracting independent frames.}
     \label{fig:amplitudeimg}
\end{figure*}
\end{comment}

\subsection{Quantitative Noise Measurement and Sensitivity}
\label{subsec:sensitivity}
Here we quantify the temporal and spatial noise of the CMOS phase camera.
The noise performance is characterized using the experimental layout described in Section~\S\ref{sec:exp_sense}. To characterize the DC power saturation levels, we vary the intensity of the 1064~nm laser beam incident on the CMOS sensor. We observe pixel saturation at $6\mu$W of incident power with beam radius of 0.25~mm.
Additional camera performance parameters are summarized in Table~\ref{tab:CameraStats}.

To determine the RF sensing capabilities and noise limitations of the phase camera the DC intensity of the beam incident on the sensor is held constant while the power modulation index is varied by sweeping the cavity, as illustrated in Fig.~\ref{fig:exp_setup}. Using this method, we report performance measurements for power modulation index values from zero to 0.046. The AC calibration factor, calculated in section~\S\ref{subsec:calibration}, is used to convert the measured noise into equivalent number of photoelectrons. The total noise in an individual pixel in a single frame is estimated by adding in quadrature photon shot noise, fixed pattern noise, and electronics noise, all of which are measured with an integration time of 32 ms.
The total amplitude and phase noise are measured as the standard deviation of the corresponding image obtained by subtracting two independent illuminated frames and dividing the result by $\sqrt{2}$.  The results in Fig.~\ref{fig:ampnoise} show a close agreement between measured and estimated noise. We find that the shot noise limit is a factor of 8 below the total measured noise and the camera sensitivity is limited by background electronic noise and fixed pattern noise.

\begin{figure}%
\center
\includegraphics[width=\linewidth]{Noise_Summary.pdf}
\caption{
  The figure above shows the phase camera noise as the power modulation index of the incident beam is varied at constant beam intensity.
  The root-mean-square amplitude noise (top) and phase noise (bottom) per pixel of the camera are calculated from a single image captured (no averaging).
  As the beam profile is the same across these measurements, the sensor area considered is the same.
  The shot noise and fixed pattern noise remain constant under these illumination conditions.
  We find that the total measured per pixel noise in phase and amplitude agrees closely with the sum of the budgeted noise sources. The current prototype of the phase camera is predominantly limited by the electronic noise and the shot noise is a factor of 8 below the total noise. The phase noise improves with higher power modulation index and with averaging of frames. The bottom plot also shows the measured per pixel phase noise with 50 frame averages in purple.
}
\label{fig:ampnoise}
\end{figure}

The demodulation pixels suppress the DC contribution of the illuminating beam using correlated balance sampling~\cite{LangeThesis, Conde:17}. However, a sufficiently high carrier field intensity will saturate the pixel of the sensor. The dynamic range is the ratio of the saturation point to the noise floor defined as~\cite{LangeThesis}
\begin{equation}
    D/R = 20~\text{log}_{10}\big(\frac{A_{sat}}{\sigma_{dark}}\big),
\end{equation}
where $A_{sat}$ is the pixel saturation value, i.e. maximum digital number and $\sigma_{dark}$ is the dark noise of the pixels. We estimate an operating dynamic range of 75~dB for the phase camera. However, due to the low modulation index of the illuminating beam in our test setup, the camera saturates in DC before reaching the full dynamic range of the amplitude and the phase readouts.

\begin{figure
\center
\includegraphics[width=\linewidth]{Frames_averaged_v2.pdf}
\includegraphics[width=\linewidth]{phase4paneimg.pdf}
\caption{Top plot: Single-frame-SNR as a function of $\Gamma_m$. The y-axis shows the single-frame-SNR, defined as SNR/$\sqrt{N_{frames}}$. The horizontal error bars show the experimental errors in the estimation of the $\Gamma_m$ using the calibrated {\it bbPD2} photo detector. We find the SNR improves with the square root of the number of frames and number of pixels, consistent with temporally and spatially independent pixel noise. Pixel averaging can be implemented to improve the SNR at the cost of spatial resolution.
The blue line represents a linear fit through the data. Using the fit, we estimate with 50  averages, the CMOS phase camera is capable of sensing RF signals in each pixel with $\Gamma_m$ as low as 0.0009. Bottom figures: Phase images -- single frame (top row) and with 50  averages (bottom row), for incident beams with low (left) and high (right) power modulation index. The phase resolution improves linearly with SNR.}
\label{fig:detectability}
\end{figure}

Camera performance for each measurement can also be quantified by calculating the signal-to-noise ratio (SNR). We define the SNR as the ratio of amplitude of the demodulated output $A$ (estimated using spatial averaging) to the measured amplitude noise $\sigma_{A}$.
\begin{equation}\label{eqn:SNR}
    SNR = \frac{A}{\sigma_{A}}
\end{equation}
We can define the SNR for individual pixels on a single frame, as well as after spatial and/or temporal averaging, Fig.~\ref{fig:detectability}.
The figure shows the SNR as a function of the power modulation index $\Gamma$ after averaging over a number of frames. The y-axis is scaled by $1/\sqrt{N_{\rm frames}}$, such that the data points line up if the noise in the frames are independent. We find the SNR scales linearly with the modulation index of the illuminating beam at constant beam intensity.
By fitting a linear model to the data, we determine the phase camera can detect signals above a modulation index of 0.0009 using 50 frame averages without any spatial averaging, see Fig.~\ref{fig:detectability}.

\begin{table*}
  	\begin{center}
  	\begin{tabular}{|p{5.5cm}|ccc|p{6cm}|}

  		\hline \hline
        \multicolumn{5}{|c|}{\textsc{Capture Settings}} \\
        \hline \hline
        Camera Parameters & ~~Minimum~~ & ~~Typical~~ & ~~Maximum~~ & Comments \\
        \hline
        Frame Rate (fps) &  -  & 7 & 60 (150) & Performance characterization was done at 7 fps. We tested the camera up to 60 fps (live-streaming limit). The camera supports up to 150 fps.  \\

        Quads & 4 & 4 & 6 & Four quad readout is measured with $\pi/2$ phase-stepped demodulation; Six quad readout is measured with $\pi/3$ phase-stepped demodulation. \\

        Sub-frames & 1 & 1 & 4 & The camera allows the capture of 1-4 sub-frames to construct a single frame. Each sub-frame is constructed with the readout of all the quads above. \\

        Quad Integration time (ms) & 1$\%$ $\text{DC}_\text{Exp}$ & 6.4 (15$\%$) & 30$\%$ $\text{DC}_\text{Exp}$ & Dependent on the frame rate, number of sub-quads and sub-frame and the readout time. The quad integration time cannot exceed 30$\%$ of the DC exposure due to the readout and dead time of the CMOS phase camera. \\

        Pixel Resolution ($\mu$m) & & 15 & & This defines the average size of each pixel on the sensor array. \\

        Sensing Area (mm) & & 4.8$\times$3.6 & & The total sensing area of the CMOS sensor. \\

        Demodulation Frequency (MHz) & 5 & 25 & 100 & The performance characterization was done at 25 MHz. The camera supports demodulation frequency up to 100 MHz. The 5 MHz lower limit is due to the RF switch. \\

        \hline \hline
        \multicolumn{5}{|c|}{\textsc{Illumination Characteristics}} \\
        \hline \hline
        DC Power (nW) [per pixel], estimated with beam size $w_0$ = 0.25~mm & 1.5 & - & 14.6 & The minimum input power for the pixels to sense the input beam and the maximum power before the pixels saturate. \\

        DC Power ($\mu$W) [sensor; for incident beam of size $w_0$ = 0.25~mm] & 0.6 & - & 6.4 & As the spot size of the incident beam increases, both the minimum and maximum DC power can be re-scaled in accordance with per pixel DC power limits above. We caution that this parameter is exposure time dependent. \\
        \hline \hline
        \multicolumn{5}{|c|}{\textsc{Frame Characterization (at 7~Hz frame rate and 15$\%$ quad integration time)}} \\
        \hline \hline
        Dynamic Range (dB) & & & 75 & The dynamic range of the camera as estimated from the dark noise level of the camera. \\

        $\Gamma_m$: Single Frame & $6.3_{-2.2}^{+2.2}~\cdot 10^{-3}$ & - & 1 & The CMOS phase camera can typically sense as low as 0.0063. \\
        SCR: Single frame & $-50_{-3}^{+3}$ dBc/pixel & - & 0 dBc/pixel & SCR down to -50~dBc/pixel can be resolved in a single frame.\\
        $\Gamma_m$: 50 frames averaged & $0.9_{-0.3}^{+0.3}$ $~\cdot 10^{-3}$  & - & 1 & With 50 frames averaged, the camera can sense incident RF beams as low as 0.0009. \\
        SCR: 50 frames averaged & $-67_{-3}^{+3}$ dBc/pixel & - & 0 dBc/pixel & With 50 frames averaged we can resolve SCR down to -67~dBc/pixel.\\

        Mean AC Noise ($e^-$) & & $10^3$ & & \\
        \hline
  	    \end{tabular}
      \end{center}
    \caption{The table summarizes the range of camera capture setting and the absolute rating for the illumination signals.}
    \label{tab:CameraStats}
\end{table*}

Using equation \ref{eqn:dBc} we can also find the camera's sensitivity limit ($SNR=1$) for resolving the  sideband-to-carrier ratio ${\rm SCR}$ in dBc per frame per pixel:
\begin{equation}\label{eqn:dBc_FRAME_PIXEL}
    {\rm SCR_{LIM}} = 20 \log_{10} \left( \frac{\Gamma_{{\rm SNR}=1}^{1{\rm fr},1{\rm pix}}}{2} \right)  = -50 \, {\rm dBc_{/fr/pix} }.
\end{equation}
Since the frame rate for the test data was $7 ~Hz$, and we only used about $50\%$ of the maximum quad integration time per frame (see Table~\ref{tab:CameraStats}) we find approximately
\begin{equation}\label{eqn:dBc_SECOND_PIXEL}
    {\rm SCR_{LIM}} = -62 \, {\rm dBc_{/sec/pix} }.
\end{equation}

\section{Signal Modeling for Point Absorbers}
\label{sec:pointAbs}
One of the critical issues limiting the sensitivity of Advanced LIGO is the presence of small (few tens of $um$ diameter) absorptive defects on the test masses, referred to as  {\it point absorbers}. Exposed to the laser field in the interferometer arms, these test mass defects cause local heating and result in local optical path length distortions for the laser field. These optical distortions excite higher-order modes in LIGO's coupled cavities, leading to excess optical loss and limiting the sensitivity of the detectors. Furthermore, the path distortions affect the carrier and sideband phase fronts differently, thus deteriorating the alignment and angular control error signals. A phase camera is capable of mapping these phase front distortions. Error signals can be extracted from the camera output and can be used to control corrective actuators. Unlike conventional quadrant photodiodes, the phase cameras offer a high spatial resolution to resolve the phase front changes due to the point absorbers.

\begin{figure}
    \centering
    \includegraphics[width=\linewidth]{Phase_cam_Point_Abs_Model.pdf}
    \caption{The figure illustrates the simplified Advanced LIGO setup for point absorber modeling. To simplify the simulation we do not include the effects of the power and the signal recycling cavities. Instead, we model the input power to be equal to the power at the beamsplitter during the Advanced LIGO O3 run~\cite{comm_O3}. The other parameters of the interferometer are summarized in the table~\ref{tab:SimParams}. The optical path distortions due to the point absorbers is modeled with a Lorentzian profile for each of the input test masses. The carrier and the sideband fields are calculated under plane beam/paraxial approximation and the corresponding beat signals are calculated at the anti-symmetric port of the beamsplitter. }
    \label{fig:PAModel}
\end{figure}

\begin{table}
  	\begin{center}
  	\begin{tabular}{|p{4cm}c|}

  		\hline \hline
        \multicolumn{2}{|c|}{\textsc{Simulation Parameters}} \\
        \hline \hline
        Differential arm offset & 10~pm \\
        Arm cavity finesse & 446 \\
        $P_{\text{Homodyne}}$ ref. Beam & 50 mW \\
        $P^{\text{BS}}_{\text{in}}$ & 1500 W \\
        $w^{\text{BS}}_{0}$ & 6~cm \\
        $L_{\text{Schnupp}}$ & 8~cm \\
        Carrier recycling gain  & 40 \\
        $45$~MHz recycling gain  & 2 \\
        $45$~MHz phase mod. index  &  \\
        \,\,\,\,\,\,\,\,at input & 0.18 \\
        \,\,\,\,\,\,\,\,at beamsplitter &0.04 \\
        \hline
  	    \end{tabular}
      \end{center}
    \caption{Simulation Parameters of the interferometer shown in Fig.~\ref{fig:PAModel}. They approximately correspond to the Advanced LIGO parameters \cite{comm_O3}.}
    \label{tab:SimParams}
\end{table}

To get an approximate estimate of the phase camera's ability to sense the effect of point absorbers LIGO's input test masses, we present a simplified model without considering the coupled cavity layout of LIGO, shown in Fig.~\ref{fig:PAModel}. We initially also ignore Gouy phase shifts because beamsplitter and input test masses are essentially in the same Gouy phase. We will get back to the effect of the output beam Gouy phase shift.
While these are oversimplifications, they still allow us the estimate the required sensitivity to pick up the point absorber phase distortions in a phase camera image taken at the interferometer anti-symmetric port.
We consider two scenarios, the current Advanced LIGO, which uses DC readout, and the A+ upgrade, which uses homodyne readout without differential arm DC offset. We choose the model parameters in accordance with the existing Advanced LIGO facilities. We assume the carrier and the 9~MHz and 45~MHz sidebands (with modulation index {$\Gamma$}) are incident at the 50/50 beamsplitter. Considering the approximate power-recycling gain of 40 for the carrier and 2 for the 45MHz sidebands we can estimate the power modulation index of the beam incident on the beamsplitter~\cite{comm_O3, craig2021}. The beam then propagates from the beamsplitter to the input test mass in each of the arm cavities. We model the response of the point absorbers to first-order, which affects the phase of the sidebands, but leaves the carrier unperturbed. The carrier experiences only a phase shift due to the DARM offset between the two arm cavities:

\begin{align}\label{eqn:iq}
r_x^{car} & = + 1 \cdot e^{i\Phi_{DARM} /2} & r_y^{car} & = + 1 \cdot e^{-i\Phi_{DARM} /2} \\
r_x^{sb} & =  - 1 \cdot e^{i 2 k \text{X} (\Vec{x},\Vec{y})} & r_y^{sb} & =  - 1 \cdot e^{i 2 k \text{Y} (\Vec{x},\Vec{y}) }
\end{align}
\noindent
where $\text{X}(\Vec{x},\Vec{y})$ and $\text{Y}(\Vec{x},\Vec{y})$ are the one-way transmission maps encoding the optical path distortions. Under these simplified assumptions one can calculate the fields of the carrier and sidebands at the anti-symmetric port of the beamsplitter. The beat map between the carrier or the reference beam with the sideband in $\mathcal{I}$ and $\mathcal{Q}$ is given by
\begin{align}
    \mathcal{I}(x,y)&=\mathbb{R}e(sb_{+}^{*} c + c^{*} sb_{-} ) \\
    \mathcal{Q}(x,y)&=\mathbb{I}m(sb_{+}^{*} c + c^{*} sb_{-} )
\end{align}

\begin{figure}
\center
\includegraphics[width=\linewidth]{PA_Model_Res_v2.pdf}
\caption{Size of the expected power modulation index in the interferometer anti-symmetric port $\mathcal{I}$-quadrature, as a function of the optical path distortions (OPD) due to a point absorber on an input test mass.
The power modulation index scales linearly with the OPD before it plateaus to a constant, which is caused due to the sidebands, 9~MHz and 45~MHz, leaking through the AS port.
The red line show the minimum power modulation index that the CMOS phase camera can sense after 50 averaged frames. In homodyne readout, where there is no large orthogonal $\mathcal{Q}$-quadrature signal to compete with, we can sense distortions greater than 0.1~nm using 50 averaged frame.
In the current DC readout there is a large $\mathcal{Q}$-quadrature signal due to the differential arm DC offset.
The phase resolution of the CMOS phase camera will thus limit sensing capabilities. The dashed black line represents the per pixel phase resolution limit after 50 frame averages (see Fig.~\ref{fig:ampnoise}).
Assume the camera is place in the appropriate Gouy phase, we can resolve optical path distortions greater than 2~nm in the DC readout scheme of current Advanced LIGO detectors. Typical optical path distortions due to these point absorbers in Advanced LIGO ranges from few nanometers to a few hundreds of nanometer ~\cite{brooks2021point, comm_O3}.
The parameters of the simulation are summarized in Table~\ref{tab:SimParams}.}
\label{fig:PA_res}
\end{figure}

One can analytically show that in the Gouy phase of the beamsplitter ($\phi=0$) the beat in $\mathcal{I}(x,y)$ and $\mathcal{Q}(x,y)$ between the carrier and the sidebands at the AS port is first-order independent of the optical path distortions due to the point absorber. We recover the usual DC readout terms:
\begin{align}\label{eq:IQzero}
    \mathcal{I}(\Vec{x},\Vec{y}) &= 0  \\
    \mathcal{Q}(\Vec{x},\Vec{y}) &= \Gamma p_{in} (\Vec{x},\Vec{y})  \sin(\frac{\Phi_{DC}}{2}) \sin(\frac{\omega_s L_s}{c})
\end{align}
However, this does not remain true at every Gouy phase in the readout beam. The small point absorber distortion evolves differently with Gouy phase. For simplicity, we can assume that the distortion fields $\text{X} (\Vec{x},\Vec{y})$ and $\text{X} (\Vec{x},\Vec{y})$ contains only one higher-order mode of order $N=l+m$. ($l$ and $k$ are for example the Hermite-Gauss mode orders.) Then the $\mathcal{I}$ readout quadrature become
\begin{align}\label{eq:Iphi}
    \mathcal{I}^{\phi} &= k \Gamma p_{in} \sin(\frac{\Phi_{DC}}{2}) \cos(\frac{\omega_s L_s}{c}) \sin(N \phi) (\text{X} - \text{Y})
\end{align}
where the spatial shape is given by the beat of the fundamental and the N-th order mode, $\Psi_N(\Vec{x},\Vec{y})\cdot \Psi_0(\Vec{x},\Vec{y})$.
Equation \ref{eq:Iphi} can be generalized by expanding $\text{X} (\Vec{x},\Vec{y})$ and $\text{X} (\Vec{x},\Vec{y})$ in terms of higher order Gaussian modes, which is straight forward, a little complicated, and not necessary if we are only interested in the camera sensitivity limitation.

The presence of a large beat signal in the $\mathcal{Q}$ quadrature from the DC readout scheme means that we also have to worry about the camera phase resolution. The beat map phase rotation is given by
\begin{align}\label{eq:IpQphi}
    \frac{\mathcal{I}}{\mathcal{Q}}^{\phi} &= k \cot(\frac{\omega_s L_s}{c}) \sin(N \phi) (\text{X} - \text{Y})
\end{align}

During the A+ upgrade Advanced LIGO will switch to a homodyne readout scheme, reducing the DC offset to zero, and instead using a separate carrier reference beam as local oscillator. We assume that the reference beam has about the same amplitude as the current carrier due to the DC offset, as this permits using the same sensing and readout electronics. Thus, the expression \ref{eq:Iphi} for sensing the effect from the point absorbers remains essentially the same, with one difference: For diagnostic purposes we now have control over the phase $\alpha_{\text{Hom}}$ of the reference beam. The $\mathcal{I}$ quadrature thus becomes
\begin{align}\label{eq:IphiHom}
    \mathcal{I}^{\phi} &= k \Gamma \sqrt{p_{ref} p_{in} } \cos(\frac{\omega_s L_s}{c}) \sin(N \phi-\alpha_{\text{Hom}}) (\text{X} - \text{Y})
\end{align}
Thus we can pick the beamsplitter Gouy phase $\phi=0$ a $90\deg$ rotated reference beam, $\alpha_{\text{Hom}}=\pi/2$, removing the large beat signal in the $\mathcal{Q}$ quadrature and avoiding the phase resolution limitation of the phase camera.

The power modulation index of the beat signal is given by
\begin{equation}\label{eq:gamma_PA}
    \Gamma_{\mathcal{I}} = \frac{2 \mathcal{I}}{DC};~~~
    \Gamma_{\mathcal{Q}} = \frac{2 \mathcal{Q}}{DC}
\end{equation}
Figure Eq.~\ref{fig:PA_res} shows the size of the expected power modulation index signal as a function of the optical path distortion (OPD). The red horizontal dashed line represents the approximate power modulation index sensitivity limit of the phase camera for each pixel. The vertical black dashed lime corresponds to the phase resolution limit of the camera, and is relevant in the presence of a large signal in the orthogonal quadrature due to the interferometer differential arm fringe offset.
We expect the phase camera to be sensitive enough to pickup optical path distortions greater than about $\sim 2~nm$ where the readout is limited by phase resolution of the phase camera. Otherwise, the phase camera is capable of sensing OPD due to point absorbers as low as $\sim 0.1~nm$. Typically, the OPD caused due to point absorbers ranges between tens up to a few hundreds of nanometers \cite{brooks2021point, comm_O3}. Currently Hartmann wavefront sensors are used to image point absorbers in aLIGO directly. These sensors map the point absorbers onto the surface of the test masses ~\cite{Brooks:07}, but do not measure their impact on the interferometer. The CMOS phase camera does not directly image the point distortions, but instead measures the change in the interferometer phase front at the AS port caused due to these point defects.

\subsection{Implications for Gravitational Wave Detectors}\label{subsec:PAImp}
The simple interferometer model presented above leads us to conclude that the CMOS phase camera developed in our group is capable of diagnosing the effect of point absorbers on the LIGO input test masses when installed at the interferometer anti-symmetric port. The exact Gouy phase of the camera will matter though, as there is no signal in the beamsplitter Gouy phase. Having a separate local oscillator reference beam, either as part of the homodyne readout or as a separate local oscillator for the phase camera, will simplify the image analysis.
Lastly, we note that the model presented above does not include the signal recycling cavity and the power recycling cavity of the Advanced LIGO detector. While the power recycling cavity only filters the beam incident on the beamsplitter, the signal recycling cavity will spatially filter the effect of the point absorber, cleaning up the mode. However, with a signal recycling mirror transmission around $32\%$, the signal recycling cavity has an extremely low finesse, preserving the distortion signal, but also making modeling rather complicated.

\section{Discussion}\label{sec:results}
We demonstrate a CMOS phase camera that is capable of imaging externally modulated RF beat signals incident on the sensor with high spatial resolution. The noise levels of the camera allow sensing of RF beat signals with a power modulation index as low as 0.0009 with 50 frame averages. The phase camera also has the capability to measure the beat signals at different frequencies and is sensitive to very low incident beam power levels. Lastly, the low latency image acquisition, design compactness, and relatively low cost of the phase camera make it suitable for numerous applications in wavefront diagnostics and sensing.

The primary application of a phase camera in gravitational-wave detectors is for diagnostic purpose, imaging any unexpected phase front distortions, such as for example those induced by point absorbers on test masses (see section \ref{sec:pointAbs}).
However the phase camera can also be useful for controlling alignment and mode-matching in an interferometer. While quadrant-photodiode and bullseye-photodiode-based schemes, with \cite{magana2019sensing} and without \cite{miller2014length} optical mode converters, have the advantage of higher signal-to-noise, they offer only four ``pixels" across the beam. In particular,
the 320$\times$240 pixel  resolution of the CMOS phase camera provides the sensing capabilities to operate and control interferometers with higher-order Laguerre-Gauss or Hermite-Gauss modes as the operating resonant mode, a scheme that was proposed to reduce the coupling of thermal noise to the gravitational readout~\cite{PhysRevLett.105.231102, carbone2013generation}.
Additionally, it has been shown that these cameras can be operated synchronously~\cite{Shrestha:2016}, which allows for multiple cameras to simultaneously record the beam in separate Gouy phases.

In summary, we expect that the CMOS phase camera will be an excellent tool for commissioning Advanced LIGO, A+, and future gravitational wave detectors such as Cosmic Explorer~\cite{CE2020LF} or Einstein Telescope~\cite{ET_design}, and might also have control applications.

\section{Acknowledgment}\label{sec:ack}
This research was funded by the National Science Foundation through the awards NSF PHY-1352511 and NSF PHY-1912536. We also would like thank numerous colleagues in the LIGO, Virgo and KAGRA scientific collaborations for many fruitful discussions.
\vspace{.5cm}

\section{Appendix}\label{sec:app}
\subsection{Imaging of Multiple Modulation Frequencies with the CMOS phase camera}\label{appsec:ff_scheme}

As discussed in section~\S\ref{sec:electronics}, a flip-flop~(FF) can be integrated with the CMOS phase camera to support four or six quad exposures. The flip-flop also allows the camera to subsequently image beat signals at different modulation frequencies. The electronic setup for this operation is shown in figure~\ref{fig:ff_design}. An RF-switch is used to switch between the modulating frequencies of the local oscillators following the frame capture sequence governed by the microcontroller. The frequency and phase stepped output of the DDS are phase-locked by ensuring the updates are triggered on the rising edge of the local-oscillator by the flip-flop. The functionality of other elements in this scheme is the same as discussed in section~\S\ref{sec:electronics}. This setup is particularly useful for application in Advanced LIGO, which allows the CMOS phase camera to image beat signals at 9~MHz and 45~MHz in real time. Using this technique, the CMOS phase camera can image beat signals at different modulation frequencies with a low-latency of 1~Hz.

\begin{figure}[h]
\center
\includegraphics[width=\linewidth]{Wiring_Diagram_Multiple.pdf}
\caption{Alternate capture scheme mentioned in section~\S\ref{sec:electronics}. It allows simultaneous imaging of RF signals with different modulation frequencies by selecting a separate modulation frequencies for every frame. In this scheme, the microcontroller controls an additional RF switch at the input off the DDS board. A 2-way splitter is used to provide a clock signal input to a flip-flop circuit. The flip-flop ensures a phase-locked output from the DDS by triggering the DDS update on the rising edge of the external local oscillator. The rest of the design block functionality is the same as discussed in Fig.~\ref{fig:signalchain}. The RF switch at the output of the DDS can be discarded if six sub-quads are desired for image capture. In this scenario, the DDS is configured to phase-step one particular channel in synchronization with the quadrature pulses received by the microcontroller.
}
\label{fig:ff_design}
\end{figure}

\bibliographystyle{apsrev4-1}

redpajama_set_name: RedPajamaArXiv


text:
namespace content {
class BrowserContext;
}

namespace extensions {
class Extension;

// ExtensionCommandsGlobalRegistry is a class that handles the cross-platform
// implementation of the global shortcut registration for the Extension Commands
// API).
// Note: It handles regular extension commands (not browserAction and pageAction
// popups, which are not bindable to global shortcuts). This class registers the
// accelerators on behalf of the extensions and routes the commands to them via
// the BrowserEventRouter.
class ExtensionCommandsGlobalRegistry
    : public BrowserContextKeyedAPI,
      public ExtensionKeybindingRegistry,
      public GlobalShortcutListener::Observer {
 public:
  // BrowserContextKeyedAPI implementation.
  static BrowserContextKeyedAPIFactory<ExtensionCommandsGlobalRegistry>*
      GetFactoryInstance();

  // Convenience method to get the ExtensionCommandsGlobalRegistry for a
  // profile.
  static ExtensionCommandsGlobalRegistry* Get(content::BrowserContext* context);

  // Enables/Disables global shortcut handling in Chrome.
  static void SetShortcutHandlingSuspended(bool suspended);

  explicit ExtensionCommandsGlobalRegistry(content::BrowserContext* context);
  virtual ~ExtensionCommandsGlobalRegistry();

  // Returns which non-global command registry is active (belonging to the
  // currently active window).
  ExtensionKeybindingRegistry* registry_for_active_window() {
    return registry_for_active_window_;
  }

  void set_registry_for_active_window(ExtensionKeybindingRegistry* registry) {
    registry_for_active_window_ = registry;
  }

  // Returns whether |accelerator| is registered on the registry for the active
  // window or on the global registry.
  bool IsRegistered(const ui::Accelerator& accelerator);

 private:
  friend class BrowserContextKeyedAPIFactory<ExtensionCommandsGlobalRegistry>;

  // BrowserContextKeyedAPI implementation.
  static const char* service_name() {
    return "ExtensionCommandsGlobalRegistry";
  }

  // Overridden from ExtensionKeybindingRegistry:
  virtual void AddExtensionKeybinding(
      const Extension* extension,
      const std::string& command_name) OVERRIDE;
  virtual void RemoveExtensionKeybindingImpl(
      const ui::Accelerator& accelerator,
      const std::string& command_name) OVERRIDE;

  // Called by the GlobalShortcutListener object when a shortcut this class has
  // registered for has been pressed.
  virtual void OnKeyPressed(const ui::Accelerator& accelerator) OVERRIDE;

  // Weak pointer to our browser context. Not owned by us.
  content::BrowserContext* browser_context_;

  // The global commands registry not only keeps track of global commands
  // registered, but also of which non-global command registry is active
  // (belonging to the currently active window). Only valid for TOOLKIT_VIEWS
  // and
  // NULL otherwise.
  ExtensionKeybindingRegistry* registry_for_active_window_;

  DISALLOW_COPY_AND_ASSIGN(ExtensionCommandsGlobalRegistry);
};

}  // namespace extensions

#endif  // CHROME_BROWSER_EXTENSIONS_EXTENSION_COMMANDS_GLOBAL_REGISTRY_H_

redpajama_set_name: RedPajamaGithub


text:
U of O Expands Full-Need Tuition Scholarship Program
University of the Ozarks is expanding its new full-need tuition scholarship program for students from Johnson County to the neighboring counties of Logan, Franklin, Madison and Newton.
Under the new program, which took effect this year for Johnson County students, the University covers the direct cost of tuition after all federal and state aid, including the U.S. Department of Education's Direct Subsidized Loans, has been applied. The full-need tuition scholarship does not cover room and board or books.
High school students from Logan, Franklin, Madison and Newton counties will be eligible for the full-need tuition scholarships starting in the 2019 Fall Semester.
"This program has been so popular and successful that we've decided to expand it to several counties surrounding Johnson County," said Reggie Hill, vice president for marketing and enrollment. "We are committed to making an Ozarks education both affordable and accessible to as many local students as possible and to help lower the debt that students are incurring."
Under the full-need tuition scholarship program, eligible students must be from the five designated Arkansas counties and have an expected family contribution (EFC) of $5,000 or less. The students must also be eligible for the Arkansas Challenge Scholarship and must file the Free Application for Federal Student Aid (FAFSA) by Feb. 15.
With a maximum federal Direct Subsidized Loan of $3,500 per year, the most an eligible Ozarks student would have in college loan debt for their tuition after four years would be $14,000. According to the Institute for College Access & Success, the average student loan debt of 2015 bachelor's degree recipients was $30,100.
Students must maintain a 2.7 grade point average and be enrolled in 15 credit hours per semester to retain the scholarship.
Approximately 100 current students hail from Johnson County, including a new incoming class of 40 this semester,
For more information about the full-need tuition scholarship, please contact the Office of Admission at 979-1227.


text:
— игровой персонаж серии игр Resident Evil, созданной компанией Capcom. Является протагонистом второй, четвёртой и шестой частей.

Во время событий RE2, Леон выступал в качестве полицейского-новичка, который с опозданием прибывает в Раккун-сити в свой первый рабочий день. Спустя шесть лет, в Resident Evil 4, Леон возвращается в качестве секретного агента, работающего на американское правительство, отправленного в Испанию на спасение дочери президента США, похищенной по пути домой из колледжа. Леон также выступает в качестве основного персонажа в ряде других игр и играет главную роль в трёх анимационных лентах — Обитель зла: Вырождение, Обитель зла: Проклятие и одну из главных в Обитель зла: Вендетта. Из игровых фильмов франшизы Леон Кеннеди появился в пятой части, Обитель зла: Возмездие, которая вышла на экраны 14 сентября 2012 года. Роль персонажа исполнил Йохан Урб.

Концепция и дизайн
Персонаж был создан Хидэки Камией как противоположность Крису Редфилду, протагонисту оригинального Resident Evil. Криса дизайнер считал «грубоватым, крутым парнем». Также Камия признался, что первоначально ему импонировал характер Криса. Однако, по мере разработки и развития нового персонажа, он решил изменить Леона. Камия сказал, что был удивлён, насколько популярным стал Леон, и похвалил его развитие в Resident Evil 4 в «круто смотрящегося парня», добавив, что он «влюбляется в него снова и снова».

Леон Кеннеди был создан для Resident Evil 2, так как разработчики хотели использовать в создаваемой игре совершенно нового персонажа, который бы не имел опыта в ужасающей ситуации, в отличие от ранее использовавшихся героев. Первоначально Леон создавался как опытный офицер полиции, но после того, как сюжет оригинальной версии Resident Evil 2 (известной сейчас как Resident Evil 1.5) был отправлен на переработку, персонаж был значительно изменён. В новой версии Леон стал полицейским-новобранцем, только поступающим на службу.

В ноябре 2002 года Леон был заявлен в качестве протагониста четвёртой части сериала. Согласно первоначально разрабатываемой версии игры с участием Леона («fog version»), он заражается вирусом «Прародитель» () и в его левой руке появляется скрытая сила. Эта версия была расширена в 2004 году. В ней персонаж использовал своё странное заболевание в борьбе против врагов. В созданном документальном фильме, геймдизайнер, объясняя концепцию персонажей, заявил, что Леон был призван «выглядеть более крепким, но также и спокойным».

Лицо персонажа в Resident Evil 4 было смоделировано Кристианом Дюррэ, уже после создания анимации игры. Во время разработки ленты Обитель Зла: Вырождение Камия сказал, что хотел бы сделать ещё одну игру с Леоном в главной роли. Продюсер Resident Evil 5 Дзюн Такэути сказал, что поклонники сериала «действительно любят» игры с участием как Леона, так и Криса, в связи с их популярностью, и в то же время, было бы «очень драматично», если бы эти два персонажа никогда не пересеклись, а серия бы закончилась. А продюсеру Resident Evil 6 Хироюки Кобаяси так понравился Леон, что он включил его в игру, так как «он занимает центральное место в истории». И несмотря на то, что игра состоит из четырёх сюжетных линий с разными персонажами, представитель Capcom назвал его «главным главным героем».

В игре Resident Evil 2 персонаж был озвучен Полом Хаддадом. В остальных частях (Resident Evil 4, Обитель зла: Вырождение и Resident Evil: The Darkside Chronicles) Леона озвучил Пол Мерсье. В японской рекламе RE2 Леона играет Брэд Ренфро.

Внешность Леону Кеннеди в Resident Evil 2 (2019) подарил модель родом из Румынии Эдуард Бадалута.

Появления

В играх
Впервые Леон появился в Resident Evil 2 как один из двух протагонистов, вместе с Клэр Редфилд. По сюжету, он полицейский-новичок, прибывший в город, где едва началась эпидемия Т-вируса, и случайно повстречавший Клэр, сражающуюся с зомби. Надеясь найти укрытие, они убегают в городской полицейский участок Раккун-сити. Поняв, что спасения там не найти, Леон и Клэр находят путь в канализацию, через которую им удаётся добраться до подземного исследовательского комплекса корпорации Umbrella, ответственной за утечку вируса. Когда Леон впервые попадает на автостоянку полицейского участка, он встречает таинственную женщину по имени Ада Вонг. В дальнейшем выяснилось, что она является шпионкой, ищущей образец G-вируса. Во время финальной битвы против Тирана модели Т-103, преследующего героев, Ада бросает вниз ракетную установку, с помощью которой Леон (либо Клэр, в зависимости от сценария) убивает его. В конце Леон сталкивается и убивает Уильяма Биркина (в его окончательной форме), и убегает вместе с Клэр и Шерри Биркин из готового взлететь на воздух комплекса.

В эпилоге Resident Evil 3: Nemesis, после прохождения игры, рассказывается, что в дальнейшем Леон стал работать на федеральное американское правительство. Во время событий игры Resident Evil Code: Veronica Клэр, застрявшая на острове Рокфор, связывается с Леоном для передачи информации её брату Крису. Сюжет рельсового шутера Resident Evil: The Darkside Chronicles для Wii включает три главы: в двух по-иному пересказываются события Resident Evil 2 и Resident Evil Code: Veronica, а третья является новой. В ней в качестве главных героев появляется Леон и Джек Краузер, которые были отправлены (в 2002 году) в Южную Америку с миссией найти Хавьера Идальго, бывшего наркобарона, который, согласно сообщениям, имел связи с корпорацией Umbrella.

Леон был главным героем Resident Evil 4. Спустя шесть лет после событий в Раккун-сити Леон, ставший специальным агентом Секретной службы, отправлен на спасение дочери американского президента Эшли Грэхэм, которую удерживают в Испании. По ходу выясняется, что похитители девушки — культ, известный как Лос Иллюминадос. Его служители взяли под контроль местных жителей при помощи паразитов, известных как Лас-Плагас. Во время поисков Эшли Леона также схватывают и вводят паразита. При помощи Луиса Серы, исследователя Лос Иллюминадос, и вернувшейся Ады Вонг, Леон удаляет паразита и спасает Эшли. В конце игры Леон с помощью Ады побеждает Озмунда Саддлера, лидера Лос Иллюминадос. Тем не менее, образец Лас-Плагас остаётся в руках убегающей на вертолёте Ады. А Леон и Эшли убегают с острова на её водном мотоцикле.

В Resident Evil 6 Леон является одним из трёх главных героев, вместе с Крисом Редфилдом и Джейком Мюллером.

Леон также появляется в нескольких неканонических играх серии. В игре Resident Evil Gaiden он является одним из главных героев, вместе с Барри Бёртоном. А в браузерной и мобильной игре Resident Evil: Zombie Busters он один из двух игровых персонажей, вместе с Клэр Редфилд. Он также появляется в игре-реинкарнации событий Раккун-сити, шутере от третьего лица Resident Evil: Operation Raccoon City. Здесь игрок управляет оперативниками корпорации Umbrella, посланными убить всех выживших. Некоторые действия в игре могут привести к смерти Леона. Леон также доступен в качестве игрового персонажа в режиме игры «Heroes Mode».

В качестве одного из главных героев вновь появился в ремейке Resident Evil 2 2019 года.

В других медиа
Леон появлялся в еженедельном маньхуа под названием Shēnghuà Wēijī 2 (), выходившем в период с 1998 года по 1999 год. А в 1999 году была выпущена романтическая комедия Èlíng Gǔbǎo II (), пересказывающая историю Resident Evil 2. В центре сюжета — Леон, Клер и Ада. Леон также является персонажем в комиксе Resident Evil (от Image Comics) и в двух новеллах —  и .

В CG-фильме «Обитель зла: Вырождение» Леон, секретный правительственный агент, вновь объединяется с Клэр Редфилд, чтобы остановить ещё одну вспышку Т-вируса. В сиквеле картины под названием «Обитель зла: Проклятие» он также принимает участие.В 2017 году выходит анимационный CG-фильм Resident evil: vendetta, в котором он является одним из главных героев. В отличие от фильмов Пола Андерсона, события этих трёх CG-лент разворачиваются в той же вселенной, что и игры, что делает их каноническими.

В интервью режиссёр Пол Андерсон сказал, что если фильм «Обитель зла: Жизнь после смерти» станет успешным, то он будет снимать пятую часть и хотел бы, чтобы Леон в ней появился. Йохан Урб, актёр, исполняющий роль Леона в кинофильме «Обитель зла: Возмездие», заявил:
Об отношениях Леона и Ады, Урб сказал: «Это наподобие отношений Малдера и Скалли в сериале "Секретные материалы", где вы ждёте, что что-то произойдёт, но этого никогда не случается. Может, в дальнейшем, я надеюсь».

В товарах
В 2004 году Capcom анонсировала серию одежды, основанную на игровых костюмах Леона. Серия была названа «Коллекцией Леона» (). В продаже также начали появляться другие товары. Так, появились две экшен-фигурки Леона от Hot Toys, три — от , и ещё несколько от других производителей, включая  , и от Capcom.,

Отзывы и критика
С момента своего появления в  Resident Evil 2, Леон получил очень позитивные отзывы. Вместе с Адой, Леон был добавлен в 2007 году ресурсом The Inquirer в список самых запоминающихся любовных команд в видеоиграх. В 2010 году Nintendo Power поставил Леона на 14-е место в своём списке самых любимых героев игр для Nintendo, отметив, что он прошёл путь от «прославившегося полицейского с плохой причёской» до крутого парня.

В 2009 году GameSpot выставлял персонажа в своём опросе на звание «All Time Greatest Game Hero». По результатам Леон обошёл ДжейСи Дентона, но проиграл Дюку Нюкему. В опросе на самого популярного персонажа видеоигры, проведённом Famitsu в 2010 году, читатели поставили Леона на 31-е место, а в книге Guinness World Records Gamer's Edition от 2011 года, ему было дано 36-е место в рейтинге лучших персонажей видеоигр. IGN включила его в список желаемых персонажей, которых бы они хотели в Resident Evil 6, называя Леона одним из двух главных героев серии вместе с Крисом Редфилдом, прозвав его «пуританским героем» сериала в статье о лучших бойцах с зомби.

GameDaily прогнозировала, что Леон вместе с Клэр Редфилд могли быть вновь героями следующего тайтла Resident Evil, ссылаясь на то, что Леона не видели со времён выхода Resident Evil 4. А один из авторов Gameplanet в своей статье заявил, что если Леон и Крис станут главными героями в очередном Resident Evil, то это будет «удивительно».

IGN неоднократно добавляла его в качестве персонажа, которого бы хотели увидеть в серии Super Smash Bros.. GamesRadar отметил дизайн Леона в Resident Evil 4 фразой:

В 2012 году Complex добавил персонажа в свой список самых «клизменных» персонажей видеоигр за «его сарказм и угрюмость в RE4», добавив, что: «…мы наслаждались игрой больше, когда Леон не говорил». PlayStation Universe написал: «Мы видели превращение Кеннеди из симпатичного, влюблённого, мокрого по самые уши щенка в способного, суперспокойного правительственного агента, сумевшего не только выжить в Раккун-сити, но и помочь спасти президентскую дочь. Неплохо для парня, которому ещё даже не исполнилось 30-ти лет».

Журнал Empire поставил Леона на 44 место в списке 50 величайших персонажей компьютерных игр.

См. также
 Список персонажей серии Resident Evil

Примечания

Персонажи Resident Evil
Вымышленные государственные детективы

redpajama_set_name: RedPajamaWikipedia


text:
Catholic Sunday Mass Readings for March 20 2022, Third Sunday of Lent, Year C – Sunday Homily
Sunday Mass Readings for March 20 2022: Third Sunday of Lent, Year C – Lectionary: 30
Sunday Mass Readings for March 20 2022, Third Sunday of Lent, Year C
Exodus 3:1-8A, 13-15
Psalms 103: 1-2, 3-4, 6-7, 8, 11
1 Corinthians 10:1-6, 10-12
Catholic Sunday Readings for March 20 2022, Third Sunday of Lent, Year C
1st Reading – Exodus 3:1-8A, 13-15
1 Moses was tending the flock of his father-in-law Jethro, the priest of Midian. Leading the flock across the desert, he came to Horeb, the mountain of God.
2 There an angel of the LORD appeared to Moses in fire flaming out of a bush. As he looked on, he was surprised to see that the bush, though on fire, was not consumed.
3 So Moses decided, "I must go over to look at this remarkable sight, and see why the bush is not burned."
4 When the LORD saw him coming over to look at it more closely, God called out to him from the bush, "Moses! Moses!" He answered, "Here I am."
5 God said, "Come no nearer! Remove the sandals from your feet, for the place where you stand is holy ground.
6 I am the God of your fathers, " he continued, "the God of Abraham, the God of Isaac, the God of Jacob." Moses hid his face, for he was afraid to look at God.
7 But the LORD said, "I have witnessed the affliction of my people in Egypt
and have heard their cry of complaint against their slave drivers, so I know well what they are suffering.
8A Therefore I have come down to rescue them from the hands of the Egyptians and lead them out of that land into a good and spacious land, a land flowing with milk and honey."
13 Moses said to God, "But when I go to the Israelites and say to them, 'The God of your fathers has sent me to you,' if they ask me, 'What is his name?' what am I to tell them?"
14 God replied, "I am who am." Then he added, "This is what you shall tell the Israelites: I AM sent me to you."
15 God spoke further to Moses, "Thus shall you say to the Israelites: The LORD, the God of your fathers, the God of Abraham, the God of Isaac, the God of Jacob, has sent me to you. "This is my name forever; thus am I to be remembered through all generations."
Responsorial Psalm – Psalms 103: 1-2, 3-4, 6-7, 8, 11.
heals all your ills,
crowns you with kindness and compassion.
6 The LORD secures justice
and the rights of all the oppressed.
7 He has made known his ways to Moses,
and his deeds to the children of Israel.
8 Merciful and gracious is the LORD,
slow to anger and abounding in kindness.
2nd Reading – 1 Corinthians 10:1-6, 10-12
1 I do not want you to be unaware, brothers and sisters, that our ancestors were all under the cloud and all passed through the sea,
2 and all of them were baptized into Moses in the cloud and in the sea.
3 All ate the same spiritual food,
4 and all drank the same spiritual drink, for they drank from a spiritual rock that followed them, and the rock was the Christ.
5 Yet God was not pleased with most of them, for they were struck down in the desert.
6 These things happened as examples for us, so that we might not desire evil things, as they did.
10 Do not grumble as some of them did, and suffered death by the destroyer.
11 These things happened to them as an example, and they have been written down as a warning to us, upon whom the end of the ages has come.
12 Therefore, whoever thinks he is standing secure should take care not to fall.
Verse Before The Gospel – Matthew 4:17
17 Repent, says the Lord;
the kingdom of heaven is at hand.
Gospel – Luke 13:1-9
1 Some people told Jesus about the Galileans whose blood Pilate had mingled with the blood of their sacrifices.
2 Jesus said to them in reply, "Do you think that because these Galileans suffered in this way they were greater sinners than all other Galileans?
3 By no means! But I tell you, if you do not repent, you will all perish as they did!
4 Or those eighteen people who were killed when the tower at Siloam fell on them – do you think they were more guilty than everyone else who lived in Jerusalem?
6 And he told them this parable: "There once was a person who had a fig tree planted in his orchard, and when he came in search of fruit on it but found none,
7 he said to the gardener, 'For three years now I have come in search of fruit on this fig tree but have found none. So cut it down. Why should it exhaust the soil?'
8 He said to him in reply, 'Sir, leave it for this year also, and I shall cultivate the ground around it and fertilize it;
9 it may bear fruit in the future. If not you can cut it down.'"
Tomorrow's Catholic Daily Readings for March 21 2022, Monday of the Third Week of Lent – Daily Homily
Sunday Mass Readings Homily – March 20 2022, Third Sunday of Lent, Year C
March 20 2022 Catholic Sunday Readings Homily Theme: The Parable of the Barren Fig Tree – Luke 13:1-9
The Parable of the Barren Fig Tree – Luke 13:1-9
Saint of the day for March 20 2022
Catholic Saint Feast Days in March
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